Thyristor-Based Optoelectronic Oscillator with Tunable Frequency and Optical Phase Lock Loop Employing Same

ABSTRACT

An optoelectronic circuit for producing an optical clock signal that includes an optical thyristor, a waveguide structure and control circuitry. The waveguide structure is configured to split an optical pulse produced by the optical thyristor such that a first portion of such optical pulse is output as part of the optical clock signal and a second portion of such optical pulse is guided back to the optical thyristor to produce another optical pulse that is output as part of the optical clock signal. The control circuitry is operably coupled to terminals of the optical thyristor and receives first and second control signal inputs. The control circuitry is configured to selectively decrease frequency of the optical clock signal based on the first control signal input and to selectively increase frequency of the optical clock signal based on the second control signal input.

BACKGROUND

1. Field

The present disclosure relates to optical communication systems and, inparticular, to receivers of optical communication signals as well asoptical and electro-optical components thereof.

2. State of the Art

Phase-shift keying (PSK) is a digital modulation scheme used in opticalcommunication that conveys data by changing, or modulating, the phase ofa reference optical signal (the optical carrier wave). PSK uses a finitenumber of phases, each assigned a unique pattern of binary digits.Usually, each phase encodes an equal number of bits. Each pattern ofbits forms the symbol that is represented by the particular phase. Thedemodulator (or receiver), which is designed specifically for thesymbol-set used by the modulator (or transmitter), determines the phaseof the received optical signal and maps it back to the symbol itrepresents, thus recovering the original data. One form of demodulationrequires that the receiver be able to compare the phase of the receivedoptical signal to an optical local oscillating signal whose phase tracksthe phase of the reference optical signal. Such a receiver is commonlyreferred to as a coherent PSK optical receiver. The coherent PSK opticalreceiver typically employs an optical phase lock loop to synchronize thephase of the optical local oscillating signal to the phase of thereference optical signal used by the transmitter.

Constellation diagrams are commonly used to represent PSK schemes. Theconstellation diagram shows points in the complex plane where, in thiscontext, the real and imaginary axes are termed the in-phase andquadrature axes respectively due to their 90° separation. Such arepresentation on perpendicular axes lends itself to straightforwardimplementation. Specifically, the constellation points are typicallyselected with uniform angular spacing around a circle. This givesmaximum phase-separation between adjacent points and thus the bestimmunity to corruption. They are positioned on a circle so that they canall be transmitted with the same energy. In this way, the moduli of thecomplex numbers they represent will be the same and the amplitude of thephase-modulated carrier wave does not change.

Two common PSK schemes are binary phase-shift keying (BPSK or 2-PSK)which uses two phases separated by 180 degrees, and quadraturephase-shift keying (QPSK or 4-PSK) or which uses four phases separatedby 90 degrees, although any number of phases may be used. Since the datato be conveyed are usually binary (or digital) in nature, the PSK schemeis usually designed with the number of constellation points being apower of 2. The higher-order PSK schemes are typically labeled with thenumber of phases used for the respective scheme. Thus, a PSK schemeemploying 8 phases separated by 45 degrees is commonly referred to as8-PSK, a PSK scheme employing 16 phases separated by 22.5 degrees iscommonly referred to 16-PSK, and a generic PSK employing M phasesseparated by (360/M) degrees is commonly referred to as M-PSK or M-aryPSK.

For the coherent BPSK optical receiver, the phase-aligned localoscillating signal can be split and mixed in the optical domain by anetwork of 3-dB fiber couplers for supply to correspondingphotodetectors as shown in FIG. 1. This configuration converts thereceived optical signal to baseband electrical signals that representsthe in-phase component and quadrature component of the received opticalsignal. These two component signals can be processed in the electricaldomain by a signal processor in order to determine the phase of thereceived optical signal and map it back to the symbol it represents,thus recovering the original data.

SUMMARY

The present disclosure relates to an optoelectronic circuit forproducing an optical clock signal that includes an optical thyristor, awaveguide structure and control circuitry. The waveguide structure isconfigured to split an optical pulse produced by the optical thyristorsuch that a first portion of such optical pulse is output as part of theoptical clock signal and a second portion of such optical pulse isguided back to the optical thyristor to produce another optical pulsethat is output as part of the optical clock signal. The controlcircuitry is operably coupled to terminals of said optical thyristor andreceives first and second control signal inputs. The control circuitryis configured to selectively decrease frequency of the optical clocksignal based on the first control signal input and to selectivelyincrease frequency of the optical clock signal based on the secondcontrol signal input.

In one embodiment, the optical thyristor can be defined by an epitaxiallayer structure that includes a bottom n-type cathode region, anintermediate p-type region formed above the bottom n-type region, anintermediate n-type region formed above the intermediate p-type region,and a top p-type anode region formed above the intermediate n-typeregion, wherein the optical thyristor further includes an anode terminalelectrically coupled to the top p-type anode region, an n-type injectorterminal electrically coupled to the intermediate n-type region, ap-type injector terminal electrically coupled to the intermediate p-typeregion, and a cathode terminal electrically coupled to the n-typecathode region.

In one embodiment, the control circuitry can include a firstfield-effect transistor whose source-drain current path is electricallycoupled between a positive voltage supply and the anode terminal. Thefirst control input signal can be supplied to the first field-effecttransistor in order to control the output power of the optical pulseproduced by the optical thyristor such that the frequency of the opticalclock signal decreases. The first field-effect transistor can be ann-channel HFET transistor having a gate terminal that receives the firstcontrol input, wherein a decrease in the first control input signalincreases the source-drain resistance of the n-channel HFET transistorsuch that the output power of the optical pulse produced by the opticalthyristor decreases and the frequency of the optical clock signaldecreases.

In one embodiment, the control circuitry can include a secondfield-effect transistor whose source-drain current path is electricallycoupled between a positive voltage supply and the n-type injectorterminal. The first control input signal can be supplied to the secondfield-effect transistor in order to control a bias current that drainselectrons from the intermediate n-type region of the optical thyristorsuch that the frequency of the optical clock signal decreases. Thesecond field-effect transistor can be a p-channel HFET transistor havinga gate terminal that receives the first control input, wherein adecrease in the first control input signal decreases the source-drainresistance of the p-channel HFET transistor such that the bias currentthat drains electrons from the intermediate n-type region of the opticalthyristor increases and the frequency of the optical clock signaldecreases.

In one embodiment, the control circuitry can include a thirdfield-effect transistor whose source-drain current path is electricallycoupled between the cathode terminal and a negative or ground voltagesupply. The second control input signal can be supplied to the thirdfield-effect transistor in order to control the output power of theoptical pulse produced by the optical thyristor such that the frequencyof the optical clock signal increases. The third field-effect transistorcan be an n-channel HFET transistor having a gate terminal that receivesthe second control input, wherein an increase in the second controlinput signal decreases the source-drain resistance of the n-channel HFETtransistor such that the output power of the optical pulse produced bythe optical thyristor increases and the frequency of the optical clocksignal increases.

In one embodiment, the control circuitry can include a fourthfield-effect transistor whose source-drain current path is electricallycoupled between the p-type injector terminal and a negative or groundvoltage supply. The second control input signal can be supplied to thesecond field-effect transistor in order to control a bias current thatdrains holes from the intermediate p-type region of the opticalthyristor such that the frequency of the optical clock signal increases.The fourth field-effect transistor can be a p-channel HFET transistorhaving a gate terminal that receives the second control input, whereinan increase in the second control input signal increases thesource-drain resistance of the p-channel HFET transistor such that thebias current that drains holes from the intermediate p-type region ofthe optical thyristor decreases and the frequency of the optical clocksignal increases.

In one embodiment, the waveguide structure can include an opticalamplifier device configured to amplify the second portion of the opticalpulse that is guided back to the optical thyristor.

In one embodiment, the intermediate n-type and p-type regions of theepitaxial layer structure of the optical thyristor include an n-typemodulation doped QW structure and a p-type modulation doped QWstructure, respectively. The control circuitry can include at last onefield effect transistor that includes an n-type QW channel formed by then-type modulation doped QW structure of the epitaxial layer structureand/or at last one field effect transistor that includes a p-type QWchannel formed by the p-type modulation doped QW structure of theepitaxial layer structure of the optical thyristor. The epitaxial layerstructure of the optical thyristor can be formed from group III-Vmaterials.

In one embodiment, the optical thyristor produces an electrical clocksignal whose frequency matches the optical clock signal output by thewaveguide structure.

In one embodiment, the optical thyristor can be realized by an opticalresonator including a closed path waveguide that supports circulatingpropagation of light and a waveguide structure that is spaced from theclosed path waveguide of the optical resonator to provide forevanescent-wave optical coupling therebetween. The waveguide structureof the optical resonator has one end disposed opposite an output end,and the optical resonator can further include a reflector structureintegral to said one end of the waveguide structure, wherein thereflector structure includes a Bragg-grating.

In another aspect, the present disclosure relates to an optical phaselock loop that includes an optoelectronic circuit as described herein,an optical phase detector, and a feedback circuit. The optoelectroniccircuit generates an optical clock signal. The optical phase detectormeasures phase difference between a reference optical signal and theoptical clock signal generated by the optoelectronic circuit. Thefeedback circuit is configured to generate first and second controlsignal inputs based on output of the optical phase detector for supplyto the optoelectronic circuit. The feedback circuit can include a chargepump circuit, which can be realized by at least one other opticalthyristor). The optical phase detector can include at least one opticalflip-flop realized by another optical thyristor. The optical phasedetector can include an optical AND gate realized by another opticalthyristor and/or an AND gate realized by a thyristor. In one embodiment,the optical phase lock loop can be configured to perform a clockrecovery function.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a schematic diagram of a prior art coherent BPSK opticalreceiver.

FIG. 2A is a schematic block diagram of coherent BPSK opticalcommunication system that can embody aspects of the present disclosure.

FIG. 2B is a schematic block diagram of an illustrative embodiment ofcoherent BPSK optical receiver according to the present disclosure.

FIG. 3A is a schematic circuit diagram of the optical mixer andthyristor-based optical BPSK detector of FIG. 2B.

FIG. 3B is a table of an exemplary mapping of phases to binary symbolsrepresented by the digital output signal produced by the optical BPSKdetector of FIG. 3A.

FIG. 3C is a constellation diagram of the exemplary mapping of phases tobinary symbols as shown in the table of FIG. 3B.

FIG. 4 is a schematic block diagram of coherent PSK opticalcommunication system that can embody aspects of the present disclosure.

FIG. 5 is a schematic block diagram of an illustrative embodiment ofcoherent PSK optical receiver according to the present disclosure.

FIG. 6A is a schematic diagram of an optical mixer and thyristor-basedoptical QPSK detector, suitable for use in the coherent PSK opticalreceiver of FIG. 5 that receives optical QPSK-modulated signals.

FIG. 6B is a table of an exemplary mapping of phases to four symbolsrepresented by the digital output signals produced by the optical QPSKdetector of FIG. 6A.

FIG. 6C is a constellation diagram of the exemplary mapping of phases tosymbols as shown in the table of FIG. 6B.

FIG. 7A is a schematic diagram of an optical mixer and thyristor-basedoptical 8-PSK detector, suitable for use in the coherent PSK opticalreceiver of FIG. 5 that receives optical 8-PSK-modulated signals.

FIG. 7B is a table of an exemplary mapping of phases to eight symbolsrepresented by the digital output signals produced by the optical 8-PSKdetector of FIG. 7A.

FIG. 7C is a constellation diagram of the exemplary mapping of phases tosymbols as shown in the table of FIG. 7B.

FIG. 8A is a schematic circuit diagram of a thyristor-based optical XORcircuit according to the present disclosure.

FIG. 8B is a table showing the XOR function provided by the optical XORcircuit of FIG. 8A.

FIGS. 9A and 9B, collectively, is a schematic diagram of the opticalmixer and thyristor-based optical 16-PSK detector, suitable for use inthe coherent PSK optical receiver of FIG. 5 that receives optical16-PSK-modulated signals.

FIG. 9C is a table of an exemplary mapping of phases to sixteen symbolsrepresented by the digital output signals produced by the optical 16-PSKdetector of FIGS. 9A and 9B.

FIG. 9D is a constellation diagram of the exemplary mapping of phases tosymbols as shown in the table of FIG. 9C.

FIGS. 10A and 10B, collectively, is a schematic diagram of the opticalmixer and thyristor-based optical PSK-QAM detector, suitable for use inthe coherent PSK optical receiver of FIG. 5 that receives opticalPSK-QAM-modulated signals.

FIG. 10C is a table of an exemplary mapping of phases and two amplitudemodulations (T1 and T2) to eight symbols represented by the digitaloutput signals produced by the optical PSK-QAM detector of FIGS. 10A and10B.

FIG. 10D is a constellation diagram of the exemplary mapping of phasesand two amplitude modulations (T1 and T2) to eight symbols as shown inthe table of FIG. 10C.

FIG. 10E is a schematic circuit diagram of part of the optical PSK-QAMdetector of FIGS. 10A and 10B for detecting the phases of the T2amplitude modulation level.

FIG. 10F is a table of an exemplary mapping of phases and two amplitudemodulations (T3 and T4) to eight symbols represented by the digitaloutput signals produced by the optical PSK-QAM detector of FIGS. 10A and10B.

FIG. 10G is a constellation diagram of the exemplary mapping of phasesand two amplitude modulations (T3 and T4) to eight symbols as shown inthe table of FIG. 10F.

FIG. 10H is a schematic circuit diagram of part of the optical PSK-QAMdetector of FIGS. 10A and 10B for detecting the phases of the T3amplitude modulation level.

FIG. 10I is a schematic circuit diagram of part of the optical PSK-QAMdetector of FIGS. 10A and 10B for detecting the phases of the T4amplitude modulation level.

FIG. 10J shows the full constellation diagram of the exemplary mappingof phases and four amplitude modulations (T1, T2, T3 and T4) to sixteensymbols that can be detected by the optical PSK-QAM detector of FIGS.10A and 10B.

FIG. 11 is a schematic block diagram of an exemplary embodiment of theoptical phase lock loop (OPPL) that functions as clock recovery circuitsuitable for use in the coherent optical receivers of FIGS. 2A and 5.

FIG. 12A is a schematic diagram of an exemplary embodiment of opticalphase detector suitable for use in the optical phase lock loop (clockrecovery circuit) of FIG. 11.

FIG. 12B is a table illustrating the logic function of the opticalflip-flop A of FIG. 12A.

FIG. 12C is a table illustrating the logic function of the opticalflip-flop B of FIG. 12A.

FIG. 12D is a table illustrating the logic function of the AND gate ofFIG. 12A.

FIG. 13A is a schematic circuit diagram of an exemplary embodiment of anoptical flip-flop, which is configured for use as the optical flip-flopA of FIG. 12A.

FIG. 13B is a schematic circuit diagram of an exemplary embodiment of anoptical flip flop, which is configured for use as the optical flip-flopB of FIG. 12A.

FIG. 14A is a schematic circuit diagram of an exemplary embodiment of anAND gate, which is configured for use as the AND gate of FIG. 12A.

FIG. 14B is a schematic circuit diagram of an exemplary embodiment of anoptical AND gate, which is configured for use as the AND gate of FIG.12A.

FIG. 15A is a schematic diagram of an exemplary embodiment of opticalcharge pump and loop filter circuit suitable for use in the opticalphase lock loop (clock recovery circuit) of FIG. 11.

FIG. 15B is a graph illustrating an example of a control signal Dgenerated by the optical charge pump and loop filter circuit of FIG.15A.

FIG. 15C is a graph illustrating an example of a control signal Cgenerated by the optical charge pump and loop filter circuit of FIG.15A.

FIG. 16 is a schematic diagram of an exemplary embodiment ofoptoelectronic oscillator suitable for use in the optical phase lockloop (clock recovery circuit) of FIG. 11.

FIG. 17 is a schematic illustration of an exemplary epitaxial layerstructure that can be used to realize the electrical devices,optoelectronic devices and optical devices of the various circuitarrangements described herein.

FIGS. 18A-18C are schematic illustrations of an exemplary embodiment ofan optical hybrid coupler that can be made utilizing the layer structureof FIG. 17.

FIGS. 19A and 19B are schematic illustrations of an exemplary embodimentof an n-channel HFET phototransistor device that can be made utilizingthe layer structure of FIG. 17.

FIGS. 20A and 20B are schematic illustrations of an exemplary embodimentof a p-channel HFET phototransistor device that can be made utilizingthe layer structure of FIG. 17.

FIGS. 21A-21F are schematic illustrations of an exemplary embodiment ofan optical thyristor and associated optoelectronic circuit elements thatcan be made utilizing the layer structure of FIG. 17 and configured foruse as an optoelectronic oscillator.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

FIG. 2A shows an optical communication system 100 employing opticalbinary phase shift keying (BPSK) signals. The system 100 includes atransmitter 112 having a laser local oscillator 114 that generates anoptical carrier signal at a predetermined wavelength. The intensity ofthe optical carrier signal output by the laser local oscillator 114oscillates at a predefined frequency. The laser local oscillator 114 canbe a distributed feedback (DFB) laser or other suitable laser emitter.The optical carrier signal output by the laser local oscillator 114 issupplied to an optical BPSK modulator 116 that modulates the phase ofthe optical carrier signal according to a supplied signal stream ofbinary bit values and a symbol clock. The symbol clock defines thetiming of the boundaries for the individual bit values of the signalstream. For a binary bit value of “1” in the signal stream, the opticalBPSK modulator 116 produces a phase-shifted optical carrier signal thathas a predefined corresponding phase offset (such as 0 degrees phaseoffset) relative to the carrier optical signal output by the laser localoscillator 114. For a binary bit value of “0” in the signal stream, theoptical BPSK modulator 116 produces a phase-shifted optical carriersignal that has a predefined corresponding phase offset (such as a 180degree or π phase offset) relative to the carrier optical signal outputby the laser local oscillator 114. The optical BPSK modulator 116 canfunction to incorporate time delay into the optical signal pathcorresponding to the required phase change (as Δφ=ωΔt). This can berealized by a ring resonator where the effective optical velocitythrough the ring resonator is controlled in some manner (such as inresponse to one or more electrical signals supplied to the ringresonator). The phase-shifted optical carrier signal produced by theoptical BPSK modulator 116 is carried by a communication medium 118(such as an optical fiber) to a receiver 120.

The receiver 120 receives the phase-shifted optical carrier signalcarried over the communication medium 118. The receiver 120 includes alaser local oscillator that is part of an optical phase lock loop (OPPL)122 that functions to recover the optical carrier signal from thereceived optical phase-shifted carrier signal. In this manner, the laserlocal oscillator of the OPPL 122 of the receiver 120 generates anoptical signal (referred to as an “optical local oscillating signal”)that oscillates at a frequency and phase that matches the opticalcarrier signal output by the laser local oscillator 114 of thetransmitter 112. The OPPL 122 estimates and compensates for differencesin oscillation frequency between the receiver's laser local oscillatorand the optical carrier signal of the transmitter 112. It also alignsthe phase of the receiver's laser local oscillator to the phase of theoptical carrier signal of the transmitter 112. The optical localoscillating signal generated by the OPPL 122 along with the receivedoptical phase-shifted carrier signal are processed by an optical BPSKcoherent demodulator 124 to identify the phase offset of the receivedoptical phased-shifted carrier signal relative to the optical localoscillating signal. This is a form of coherent homodyne detection of thephase information carried in the received phased-shifted optical carriersignal. The optical BPSK coherent demodulator 124 generates anelectrical signal that encodes the binary bit values corresponding tothe phase offset information of the received phased-shifted opticalcarrier signal. The correspondence between the phase offset informationand the binary bit values is dictated by the BPSK modulation schemeemployed by the optical BPSK modulator 116 of the transmitter 112. Inthis manner, the optical BPSK coherent demodulator 124 recovers thesignal stream of binary bit values carried by the received phase-shiftedoptical carrier signal. The optical BPSK coherent demodulator 124 alsooperates to generate an electrical clock signal (timing information)corresponding to the symbol clock of the transmitter 112. The recoveredsignal stream and the clock signal are supplied to circuitry 126 thatdecodes the signal stream (if necessary) and performs serial-to-paralleldata conversion. The parallel data (typically, byte-sizes chunks) can beoutput to a data processing circuit as needed.

FIG. 2B is a block diagram illustrating an embodiment of the receiver120 of FIG. 2A, which includes an OPPL 122, an optical mixer andthyristor-based optical BPSK detector 124A, an optical clock recoverblock 124B, and signal decode circuitry 126. The OPPL 122 includes alaser local oscillator 201 and a loop filter 203. The loop filter 203functions to estimate and compensate for differences in oscillationfrequency between the optical local oscillating signal generated by thelaser local oscillator 201 and the optical carrier signal of thetransmitter 112. The loop filter 203 also functions to align the phaseof the optical local oscillating signal to the phase of the opticalcarrier signal of the transmitter 112. The output of the laser localoscillator is supplied to a polarization control module 205 that fixesthe polarization state of the optical local oscillating signal to matchthe polarization of the received optical signal. This may be necessaryas the laser local oscillator 201 may produce an optical polarizationstate that differs from the received optical signal.

The optical mixer and thyristor-based optical BPSK detector 124A isconfigured to identify the phase offset of the received opticalphased-shifted carrier signal relative to the optical local oscillatingsignal. This is form of coherent homodyne detection of the phaseinformation carried in the received phased-shifted optical carriersignal. The optical mixer and thyristor-based optical BPSK detector 124Agenerates an electrical signal that encodes the binary bit valuescorresponding to the phase offset information of the receivedphased-shifted optical carrier signal. The correspondence between thephase offset information and the binary bit values is dictated by theBPSK modulation scheme employed by the optical BPSK modulator 116 of thetransmitter 112. In this manner, the optical mixer and thyristor-basedoptical BPSK detector 124A recovers the signal stream of binary bitvalues carried by the received phase-shifted optical carrier signal.

The optical clock recovery block 124B operates to generate an electricalclock signal whose ON/OFF transitions are synchronized to the symbolclock of the transmitter. This symbol clock is embedded in the opticalsignal generated by the optical mixer and thyristor-based optical BPSKdetector 124A, which represents the recovered signal stream of binarybit values carried by the received phase-shifted optical carrier signaland detected by the optical mixer and thyristor-based optical BPSKdetector 124A.

The signal decode circuitry 126 utilizes the electrical clock signal(recovered symbol clock) generated by the clock recovery block 124B tosample the pulses of the electrical signal generated by the opticalmixer and thyristor-based optical BPSK detector 124A. This electricalsignal represents the recovered signal stream of binary bit valuescarried by the received phase-shifted optical carrier signal anddetected by the optical mixer and thyristor-based optical BPSK detector124A. This sampling is performed at the timing that correspondsrecovered symbol clock and thus is aligned to the binary bit valuescarried by the received phase-shifted optical carrier signal. The signaldecode circuitry 126 also performs signal decoding operations that mapeach recovered phase sample back to the binary symbol it represents andthus recovers the original data.

An embodiment of the optical mixer and thyristor-based optical BPSKdetector 124A is shown in FIG. 3A. It employs a network of three180-degree optical hybrid couplers 300A, 300B, 300C for optical mixingof the received phase-shifted carrier signal and the optical localoscillating signal output by the OPPL 122.

Each 180-degree optical hybrid coupler includes two input waveguides(In0/In1) and two output waveguides (Out0/Out1). The two inputwaveguides In0, In1 receive two separate input optical signals S0 andS1. The 180-degree optical hybrid coupler mixes the two input opticalsignals S0 and S1 to produce two output signals that propagate out fromthe Out0 and Out1 waveguides where the power of the two input opticalsignals S0, S1 is split evenly (50:50 split for each input signal) ineach one of two output signals. Specifically, the intensity I0 of theoptical signal propagating out from the output waveguide Out0 is givenas:

I0=E _(S1) ² cos²(κL _(eff))+E _(S0) ² sin²(κL _(eff))+E _(S1) E _(S0)sin(2κL _(eff))sin φ  (1)

-   -   where E_(S1) is the field amplitude of the input optical signal        S1 supplied to the In1 input waveguide,    -   E_(S0) is the field amplitude of the input optical signal S0        supplied to the In0 input waveguide,    -   κ and L_(eff) is the coupling constant and the effective        coupling length of the 180-degree optical hybrid coupler, and    -   φ is the phase difference in the optical signals produced in the        Out0 and Out1 output waveguides.        Similarly, the intensity I1 of the optical signal propagating        out from the output waveguide Out2 is given as:

I1=E _(S1) ² sin²(κL _(eff))+E _(S0) ² cos²(κL _(eff))+E _(S1) E _(S0)sin(2κL _(eff))sin φ.  (2)

In the case of the 50:50 split ratio, κL_(eff) is π/4 and Eqn. (1) andEqn. (2) becomes:

I0=½E _(S1) ²+½E _(S0) ² +E _(S1) E _(S0) sin φ, and  (3)

I1=½E _(S1) ²+½E _(S0) ² −E _(S1) E _(S0) sin φ.  (4)

These equations show that the maximum intensity of the two outputsignals have a phase difference of 180 degrees or π radians. Moreover,there is a phase delay (lag) of 90 degrees of π/2 radians with respectto the S0 optical signal component that is part of the output signalpropagating from the output waveguide Out1 relative to the S0 opticalsignal component that is part of the output signal propagating from theoutput waveguide Out0.

As shown in FIG. 3A, the three 180-degree optical hybrid couplers (300A,300B, 300C) provide mixing of the received phase-shifted optical carriersignal and the optical LO signal output from the OPPL 122. Twowaveguides 301A, 301B supply the received phased-shifted optical carriersignal and the optical LO signal to the In0 and the In1 inputwaveguides, respectively, of the first 180-degree optical hybrid coupler300A. In this configuration, the first 180-degree optical hybrid coupler300A mixes the received phase-shifted optical carrier signal and theoptical LO signal to produce two output signals that propagate out fromthe Out0 and Out1 waveguides of the 180-degree optical hybrid coupler300A. The intensity of the two input optical signals is split evenly(50:50 split) in each one of two output signals with the maximumintensity of the two output signals having a phase difference of 180degrees (π radians).

A waveguide 303A supplies the output signal from the Out0 outputwaveguide of the 180-degree optical hybrid coupler 300A to the In0 inputwaveguide of the 180-degree optical hybrid coupler 300B. In thisconfiguration, the 180-degree optical hybrid coupler 300B produces twooutput signals that propagate out from the Out0 and the Out1 outputwaveguides of the 180-degree optical hybrid coupler 300B. The intensityof the input signal is split evenly (50:50 split) in each one of twooutput signals with the maximum intensity of the two output signalshaving a phase difference of 180 degrees (π radians).

A waveguide 303B supplies the output signal output from the Out1 outputwaveguide of the 180-degree optical hybrid coupler 300A to the In0 inputwaveguide of 180-degree optical hybrid coupler 300C. In thisconfiguration, the 180-degree optical hybrid coupler 300C produces twooutput signals that propagate out from the Out0 and Out1 outputwaveguides of the 180-degree hybrid coupler 300C. The intensity of theinput signal is split evenly (50:50 split) in each one of two outputsignals with the maximum intensity of the two output signals having aphase difference of 180 degrees (π radians).

Waveguides 305A, 305B, 305C, 305D supply the optical signals output fromthe Out0 and Out1 output waveguides of the 180-degree hybrid coupler300B as well as the optical signals output from the Out0 and Out1 outputwaveguides of the 180-degree hybrid coupler 300C to correspondingphototransistors Q1, Q2, Q3, Q4 of the thyristor-based optical phasedetector as described below.

The thyristor-based optical phase detector includes a four terminalvertical thyristor (N region-P region-N region-P region) 307 with asplit load. A first load element 309A is coupled between the positivevoltage supply (V_(cc)) and the anode terminal of the thyristor 307. Asecond load element 309B is coupled between the cathode terminal of thethyristor 307 and the negative voltage supply (ground). The thyristor307 also has a p-channel injector terminal and an n-channel injectorterminal.

A p-channel HFET phototransistor Q1 and an n-channel HFETphototransistor Q2 are coupled in series between the positive voltagesupply (V_(cc)) and the negative voltage supply (ground). The p-channelHFET phototransistor Q1 has a source and gate terminal both connected tothe positive voltage supply (V_(cc)). The n-channel HFET phototransistorQ2 has a source and gate terminal both connected to the negative voltagesupply (ground). The drain of the p-channel HFET phototransistor Q1 andthe drain of the n-channel HFET phototransistor Q2 are coupled togetherand to the p-channel injector terminal of the thyristor 307.

A p-channel HFET phototransistor Q3 and an n-channel HFETphototransistor Q4 are coupled in series between the positive voltagesupply (V_(cc)) and the negative voltage supply (ground). The p-channelHFET phototransistor Q3 has a source and gate terminal both connected tothe positive voltage supply (V_(cc)). The n-channel HFET phototransistorQ4 has a source and gate terminal both connected to the negative voltagesupply (ground). The drain of the p-channel HFET phototransistor Q3 andthe drain of the n-channel HFET phototransistor Q4 are coupled togetherand to the n-channel injector terminal of the thyristor 307.

The optical signal output from the Out0 output waveguide of the180-degree optical hybrid coupler 300B is guided into the waveguideregion of the p-channel HFET phototransistor Q1 and the optical signaloutput from the Out1 output waveguide of the 180-degree optical hybridcoupler is guided into the waveguide region of the p-channel HFETphototransistor Q3. The optical signal output from the Out0 outputwaveguide of the 180-degree optical hybrid coupler 300C is guided intothe waveguide region of the n-channel HFET phototransistor Q2 and theoptical signal output from the Out1 output waveguide of the 180-degreeoptical hybrid coupler 300C is guided into the waveguide region of then-channel HFET phototransistor Q4.

In this configuration, the phase of the optical carrier signal componentthat is part of the output signal propagating from the Out0 outputwaveguide of the 180-degree optical hybrid coupler 300B (and supplied tothe waveguide region of the p-channel HFET phototransistor Q1) isaligned (lag near 0 degrees or radians) to the phase of the receivedphase-shifted optical carrier signal. The optical carrier signalcomponent that is part of the output signal propagating from the Out1output waveguide of the 180-degree optical hybrid coupler 300B (andsupplied to the waveguide region of the p-channel HFET phototransistorQ3) lags in phase by 90 degrees (π/2 radians) the phase of the receivedphase-shifted optical carrier signal. The phase of the optical carriersignal component that is part of the output signal propagating from theOut0 output waveguide of the 180-degree optical hybrid coupler 300C (andsupplied to the waveguide region of the n-channel HFET phototransistorQ2) lags in phase by 90 degrees (π/2 radians) the phase of the receivedphase-shifted optical carrier signal. The phase of the optical carriersignal component that is part of the output signal propagating from theOut1 output waveguide of the 180-degree optical hybrid coupler 300C (andsupplied to the waveguide region of the n-channel HFET phototransistorQ4) lags in phase by 180 degrees (π radians) the phase of the receivedphase-shifted optical carrier signal.

Furthermore, in this configuration, the p-channel HFET phototransistorQ1 and the n-channel HFET phototransistor Q4 are activated and operateas turn-n phototransistors for the thyristor 307 when the phase of thereceived phase-shifted optical carrier signal corresponds to a firstpredefined phase offset (such as 0 degrees or radians). When activated,the p-channel HFET phototransistor Q1 supplies hole current to thep-channel injector and the n-channel HFET phototransistor supplieselectron current to the n-channel injector such that the thyristor 307operates in its ON state where it conducts current vertically betweenthe anode and the cathode terminal of the thyristor 307. The p-channelHFET phototransistor Q3 and the n-channel HFET phototransistor Q2 areactivated and operate as turn-off F phototransistors when the phase ofthe received phase-shifted optical carrier signal corresponds to asecond predefined phase offset (such as 180 degrees or π radians). Whenactivated, the p-channel HFET phototransistor Q3 draws electron currentfrom the n-channel injector and the n-channel HFET phototransistor Q2draws hole current from the p-channel injector such that the thyristor307 operates in its OFF state where there is minimal conductionvertically between the anode and the cathode terminal of the thyristor307.

The ON and OFF states of the thyristor 307 generates digital (binary)electrical signals A, Ā at the cathode terminal and the anode terminal,respectively. Note that the digital electrical signal A generated at thecathode terminal is complementary to the digital electrical signal Āgenerated at the anode terminal. The ON state of the thyristor 307 canalso be configured above the lasing threshold such that the thyristor307 emits a digital (ON/OFF) optical signal corresponding to the digitalelectrical signal A generated at the cathode terminal of the thyristor307. In this configuration, the binary levels of the digital electricalsignals A, Ā (and possibly the ON/OFF state of the digital opticalsignal emitted from the thyristor) for the ON and OFF states of thethyristor 307 correspond to the two possible phase offsets of thereceived phase-shifted optical carrier signal. An example ofcorrespondence between the binary levels of the digital electricalsignals A, Ā (and possibly the digital optical signal emitted by thethyristor) and two phase offsets of the received phase-shifted opticalcarrier signal is shown in the chart of FIG. 3B and the phaseconstellation diagram of FIG. 3C. Note that these two possible phaseoffsets are 180 degrees (or π radians) apart from one another.

QPSK and Higher Communication Systems

FIG. 4 shows an optical communication system 100′ employing optical QPSK(or higher order) phase shift keying signals. The transmitter 112′ issimilar to the transmitter 112 for the BPSK system with the addition ofa symbol encoder 117 that maps two sequential bits of the signal streaminto a corresponding symbol phase. The optical PSK phase modulator 116′modulates the phase of the optical carrier signal generated by the laserlocal oscillator 116 according to the symbol phase generated by thesymbol encoder 117. The optical PSK modulator 116′ can function toincorporate time delay into the optical signal path corresponding to therequired phase change (as Δφ=ωΔt). This can be realized by a ringresonator where the effective optical velocity through the ringresonator is controlled in some manner (such as in response to one ormore electrical signals supplied to the ring resonator). Thephase-shifted optical carrier signal produced by the optical PSKmodulator 116′ is carried by the communication medium (such as anoptical fiber) to the receiver 120′.

The receiver 120′ of the optical communication system 100′ is similar tothe receiver 120 of the BPSK system including an OPPL 122 that functionsto estimate and compensate for differences in oscillation frequencybetween the receiver's laser local oscillator and the optical carriersignal of the transmitter 112′. It also aligns the phase of thereceiver's laser local oscillator to the phase of the optical carriersignal of the transmitter 112′. The optical local oscillating signalgenerated by the OPPL 122 along with the received optical phase-shiftedcarrier signal are processed by the optical PSK coherent demodulator124′ to identify the phase offset of the received optical phased-shiftedcarrier signal relative to the optical local oscillating signal. This isa form of coherent homodyne detection of the phase information carriedin the received phased-shifted optical carrier signal. The optical PSKcoherent demodulator 124′ generates an electrical signal that encodesthe symbol phase data corresponding to the phase offset information ofthe received phased-shifted optical carrier signal. The correspondencebetween the phase offset information and the symbol data values isdictated by the PSK modulation scheme employed by the optical PSKmodulator 116′ of the transmitter 112′. The optical PSK coherentdemodulator 124′ also operates to generate an electrical clock signal(timing information) corresponding to the symbol clock of thetransmitter 112′. In this manner, the optical PSK coherent demodulator124′ recovers the symbol phase data carried by the receivedphase-shifted optical carrier signal. The recovered phase data and theclock signal are supplied to a phase symbol decoder 128 that decodes therecovered phase symbol data to generate the binary signal streamrepresented by the symbol data. The binary signal stream is output bythe symbol decoder 128 to circuitry 126′ that decodes the signal stream(if necessary) and performs serial-to-parallel data conversion. Theparallel data (typically, byte-sizes chunks) can be output to a dataprocessing circuit as needed.

FIG. 5 is a block diagram illustrating an embodiment of the receiver120′ of FIG. 4, which includes an OPPL 122, an optical mixer andthyristor-based optical PSK detector 124A′, an optical clock recoverblock 124B′, a symbol decoder 128 and signal decode circuitry 126. TheOPPL 122 includes a laser local oscillator 201 and a loop filter 203.The loop filter 203 functions to estimate and compensate for differencesin oscillation frequency between the optical local oscillating signalgenerated by the laser local oscillator 201 and the optical carriersignal of the transmitter 112′. The loop filter 203 also functions toalign the phase of the optical local oscillating signal to the phase ofthe optical carrier signal of the transmitter 10. The output of thelaser local oscillator is supplied to a polarization control module 205that is supplied to a polarization control module 205 that fixes thepolarization state of the optical local oscillating signal to match thepolarization of the received optical signal. This may be necessary asthe laser local oscillator 201 may produce an optical polarization statethat differs from the received optical signal.

The optical mixer and thyristor-based optical PSK detector 124A′ isconfigured to identify the phase offset of the received opticalphased-shifted carrier signal relative to the optical local oscillatingsignal. This is form of coherent homodyne detection of the phaseinformation carried in the received phased-shifted optical carriersignal. The optical mixer and thyristor-based optical PSK detector 124A′generates an electrical signal that encodes the symbol datacorresponding to the phase offset information of the receivedphased-shifted optical carrier signal. The correspondence between thephase offset information and the symbol data is dictated by the QPSK (orhigher order) modulation scheme employed by the optical PSK modulator116′ of the transmitter 112′. In this manner, the optical mixer andthyristor-based optical PSK detector 124A′ recovers the signal stream ofsymbols carried by the received phase-shifted optical carrier signal.

The optical clock recovery block 124B′ operates to generate anelectrical clock signal whose ON/OFF transitions are synchronized to thesymbol clock of the transmitter 112′. This symbol clock is embedded inthe optical signal generated by the optical mixer and thyristor-basedoptical PSK detector 124A′, which represents the recovered signal streamof symbols carried by the received phase-shifted optical carrier signaland detected by the optical mixer and thyristor-based optical PSKdetector 124A′.

The signal decode circuitry 126 utilizes the electrical clock signal(recovered symbol clock) generated by the clock recovery block 124B′ tosample the symbol data generated by the optical mixer andthyristor-based optical PSK detector 124A′. This symbol data representsthe recovered signal stream of symbols carried by the receivedphase-shifted optical carrier signal and detected by the optical mixerand thyristor-based optical PSK detector 124A′. This sampling isperformed at the timing that corresponds recovered symbol clock and thusis aligned to the binary bit values carried by the receivedphase-shifted optical carrier signal. The signal decode circuitry 126also performs signal decoding operations that map each recovered phasesample back to the binary symbol it represents and thus recovers theoriginal data.

Optical QPSK Detector

An embodiment of the optical mixer and optical PSK detector 124A′ fordetecting and demodulating optical QPSK signals is shown in FIG. 6A. Itemploys a network of two 180-degree optical hybrid couplers 600A and600B and two optical BPSK detectors 124A-1 and 124A-2 as described abovewith respect to FIG. 3A for optical mixing and demodulation of thereceived phase-shifted carrier signal and the optical local oscillatingsignal output by the OPPL 122. As shown in FIG. 6A, one 180-degreeoptical hybrid coupler 600A is used to split the received phase-shiftedcarrier signal for supply to the two optical BPSK detectors 124A-1 and124A-2. The other 180-degree optical hybrid coupler 600B is used tosplit the optical local oscillating signal output by the OPPL 122 forsupply to the two optical BPSK detectors 124A-1 and 124A-2.

Waveguide 601A supplies the received phased-shifted optical carriersignal to the In0 input waveguide of the 180-degree optical hybridcoupler 600A. In this configuration, the 180-degree optical hybridcoupler 600A produces two output signals that propagate out from theOut0 and Out1 waveguides of the 180-degree optical hybrid coupler 600A.The intensity of the received phased-shifted optical carrier signal issplit evenly (50:50 split) in each one of two output signals with aphase delay (lag) of 90 degrees (π/2 radians) for the output signal thatpropagates out from the Out1 waveguide relative the output signal thatpropagates out from the Out0 waveguide.

Waveguide 601B supplies the optical LO signal to the In0 input waveguideof the 180-degree optical hybrid coupler 600B. In this configuration,the 180-degree optical hybrid coupler 600B produces two output signalsthat propagate out from the Out0 and Out1 waveguides of the 180-degreeoptical hybrid coupler 600B. The intensity of the optical LO signal issplit evenly (50:50 split) in each one of two output signals with aphase delay (lag) of 90 degrees (π/2 radians) for the output signal thatpropagates out from the Out1 waveguide relative the output signal thatpropagates out from the Out0 waveguide.

Waveguides 603A and 603C supply the optical signals output from the Out0output waveguides of the 180-degree hybrid couplers 600A and 600B to theinput waveguides 301A and 301B of the optical BPSK detector 124A-1 asshown. Waveguides 603B and 603D supply the optical signals output fromthe Out1 output waveguides of the 180-degree hybrid couplers 600A and600B to the input waveguides 301A and 301B of optical BPSK detector124A-2 as shown. In this configuration, the phase of the receivedphase-shifted optical carrier signal that is supplied via the waveguide603A to the input waveguide 301A of the optical BPSK detector 124A-1 isaligned in phase (lag near 0 degrees or radians) with the phase of thereceived phase-shifted optical carrier signal. The phase of the receivedphase-shifted optical carrier signal that is supplied via the waveguide603B to the input waveguide 301A of the optical BPSK detector 124A-2lags by 90 degrees (π/2 radians) the phase of the received phase-shiftedoptical carrier signal. The phase of the optical LO signal that issupplied via the waveguide 603C to the input waveguide 301B of theoptical BPSK detector 124A-1 is aligned in phase (lag near zero degreesor radians) the phase of the optical LO signal produced by the OPPL 122.The phase of the optical LO signal that is supplied via the waveguide603D to the input waveguide 301B of the optical BPSK detector 124A-2lags by 90 degrees (π/2 radians) the phase of the optical LO signalproduced by the OPPL 122.

The ON and OFF states of the thyristor 307 of the optical BPSK detector124-1 generates digital electrical signals A, Ā at the cathode terminaland the anode terminal, respectively. Note that the digital electricalsignal A generated at the cathode terminal is complementary to thedigital electrical signal Ā generated at the anode terminal. The ONstate of such thyristor 307 can also be configured above the lasingthreshold such that the thyristor 307 emits a digital (ON/OFF) opticalsignal corresponding to the binary levels of the digital electricalsignal A generated at the cathode terminal of the thyristor 307 of theoptical BPSK detector 124-1. The ON and OFF states of the thyristor 307of the thyristor-based optical BPSK detector 124-2 generates digitalelectrical signals B, B at the cathode terminal and the anode terminal,respectively. Note that the digital electrical signal B generated at thecathode terminal is complementary to the digital electrical signal Bgenerated at the anode terminal. The ON state of such thyristor 307 canalso be configured above the lasing threshold such that the thyristor307 emits a digital (ON/OFF) optical signal corresponding to the digitalelectrical signal B generated at the cathode terminal of the thyristor307 of the optical BPSK detector 124-2.

In this configuration, the binary levels of the digital electricalsignals A, Ā, B, B (and possibly the ON/OFF state of the digital opticalsignals emitted from the thyristors) for the ON and OFF states of thethyristors within the optical BPSK detectors 124-1 and 124-2 correspondto the four possible phase offsets of the received phase-shifted opticalcarrier signal. An example of correspondence between the binary levelsof the digital electrical signals A, B (and possibly the ON/OFF state ofthe corresponding digital optical signals emitted from the thyristors)and four phase offsets of the received phase-shifted optical carriersignal is shown in the chart of FIG. 6B and the phase constellationdiagram of 7C. Note that these two possible phase offsets are 90 degrees(or π/2 radians) apart from one another.

Optical 8-PSK Detector

An embodiment of the optical mixer and optical PSK detector 124A″ fordetecting and demodulating optical 8-PSK signals is shown in FIG. 7A. Itemploys a network of two 180-degree optical hybrid couplers 700A and700B and two optical QPSK detectors 124A′-1 and 124A′-2 as describedabove with respect to FIG. 6A for optical mixing of the receivedphase-shifted carrier signal and the optical local oscillating signaloutput by the OPPL 122 and demodulation. As shown in FIG. 7A, the180-degree optical hybrid coupler 700A functions to split the receivedphase-shifted carrier signal for supply to the two optical QPSKdetectors 124A′-1 and 124A′-2. The 180-degree optical hybrid coupler700B functions to split the optical local oscillating signal output bythe OPPL 122 for supply to the two optical QPSK detectors 124A′-1 and124A′-2.

Waveguide 701A supplies the received phased-shifted optical carriersignal to the In0 input waveguide of the 180-degree optical hybridcoupler 700A. In this configuration, the 180-degree optical hybridcoupler 700A produces two output signals that propagate out from theOut0 and Out1 waveguides of the 180-degree optical hybrid coupler 700A.The intensity of the received phased-shifted optical carrier signal issplit evenly (50:50 split) in each one of two output signals with aphase delay (lag) of 90 degrees (π/2 radians) for the output signal thatpropagates out from the Out1 waveguide relative the output signal thatpropagates out from the Out0 waveguide.

Waveguide 701B supplies the optical LO signal to the In0 input waveguideof the 180-degree optical hybrid coupler 700B. In this configuration,the 180-degree optical hybrid coupler 700B produces two output signalsthat propagate out from the Out0 and Out1 waveguides of the 180-degreeoptical hybrid coupler 700B. The intensity of the optical LO signal issplit evenly (50:50 split) in each one of two output signals with aphase delay (lag) of 90 degrees (π/2 radians) for the output signal thatpropagates out from the Out1 waveguide relative the output signal thatpropagates out from the Out0 waveguide.

Waveguide 703A supplies the optical signal output from the Out0 outputwaveguide of the 180-degree hybrid coupler 700A to an optical phasemodulator 705, which functions to delay the phase of the suppliedreceived phased-shifted optical carrier signal by 45 degrees (or π/4radians). The resultant optical signal is supplied to the inputwaveguide 601A of the optical QPSK detector 124A′-1 as shown. Waveguide703B supplies the optical signal output from the Out1 output waveguideof the 180-degree hybrid coupler 700A to the input waveguide 601A of theoptical QPSK detector 124A′-2 as shown. Waveguide 703C supplies theoptical signal output from the Out0 output waveguide of the 180-degreehybrid coupler 700B to the input waveguide 601B of the optical QPSKdetector 124A′-1 as shown. Waveguide 703D supplies the optical signaloutput from the Out1 output waveguide of the 180-degree hybrid coupler700B to the input waveguide 601B of the optical QPSK detectors 124A′-2as shown.

In this configuration, the phase of the received phase-shifted opticalcarrier signal that is supplied via the waveguide 703A to the inputwaveguide 601A of the optical QPSK detector 124A′-1 lags by 45 degrees(π/4 radians) the phase of the received phase-shifted optical carriersignal. The phase of the received phase-shifted optical carrier signalthat is supplied via the waveguide 703B to the input waveguide 601A ofthe optical QPSK detector 124A′-2 lags by 90 degrees (π/2 radians) thephase of the received phase-shifted optical carrier signal. The phase ofthe optical LO signal that is supplied via the waveguide 703C to theinput waveguide 601B of the optical QPSK detector 124A′-1 is aligned(lag near zero degrees or radians) to the phase of the optical LO signalsupplied by the OPLL 122. The phase of the optical LO signal that issupplied via the waveguide 703D to the input waveguide 601B of opticalQPSK detector 124A′-2 lags by 90 degrees (π/2 radians) the phase of theoptical LO signal that is supplied by the OPLL 122.

Furthermore, in this configuration, the ON and OFF states of thethyristors of the optical QPSK detector 124A′-1 generates digitalelectrical signals A, Ā, B, B (and possibly corresponding digitaloptical signals A, B), and the ON and OFF states of the thyristors ofthe optical QPSK detector 124A′-2 generates digital electrical signalsA′, Ā′, B′, B′ (and possibly corresponding digital optical signals A′,B′). The digital optical signals A′, B′ are supplied to athyristor-based optical XOR block 707 that generates a digitalelectrical signal C that is the Boolean XOR function of the ON/OFFstates of the digital optical signals A′, B′. The binary levels of thedigital electrical signals A, B, C correspond to the eight possiblephase offsets of the received phase-shifted optical carrier signal. Anexample of correspondence between the binary levels of the electricalsignals A, B, C and eight phase offsets of the received phase-shiftedoptical carrier signal is shown in the chart of FIG. 7B and the phaseconstellation diagram of FIG. 7C. Note that these eight possible phaseoffsets are 45 degrees (or π/4 radians) apart from one another.

Thyristor-Based Optical XOR Block

An embodiment of the thyristor-based optical XOR block 707 is shown inFIG. 8A. It employs a network of two 180-degree optical hybrid couplers800A, 800B for splitting the input digital optical signals A′, B′. Awaveguide 801A supplies the input digital optical signal A′ to the In0input waveguide of 180-degree optical hybrid coupler 800A. In thisconfiguration, the 180-degree optical hybrid coupler 800A produces twooutput signals that propagate out from the Out0 and the Out1 outputwaveguides of the 180-degree optical hybrid coupler 800A. The intensityof the digital optical signal A′ is split evenly (50:50 split) in eachone of two output signals. A waveguide 801B supplies the input digitaloptical signal B′ to the In0 input waveguide of 180-degree opticalhybrid coupler 800B. In this configuration, the 180-degree opticalhybrid coupler 800B produces two output signals that propagate out fromthe Out0 and the Out1 output waveguides of the 180-degree optical hybridcoupler 800B. The intensity of the digital optical signal B′ is splitevenly (50:50 split) in each one of two output signals.

Waveguides 803A, 803B, 803C, 803D supply the optical signals output fromthe Out0 and Out1 output waveguides of the 180-degree hybrid couplers800A and 800B to corresponding phototransistors Q1, Q2, Q3, Q4 of athyristor-based circuit as described below.

The thyristor-based circuit includes a four terminal vertical thyristor(N region-P region-N region-P region) 807 with a split load. A firstload element 809A is coupled between the positive voltage supply(V_(cc)) and the anode terminal of the thyristor 807. A second loadelement 809B is coupled between the cathode terminal of the thyristor807 and the negative voltage supply (ground). The thyristor 807 also hasa p-channel injector terminal and an n-channel injector terminal.

A p-channel HFET phototransistor Q1 and an n-channel HFETphototransistor Q2 are coupled in series between the positive voltagesupply (V_(cc)) and the negative voltage supply (ground). The p-channelHFET phototransistor Q1 has a source and gate terminal both connected tothe positive voltage supply (V_(cc)). The n-channel HFET phototransistorQ2 has a source and gate terminal both connected to the negative voltagesupply (ground). The drain of the p-channel HFET phototransistor Q1 andthe drain of the n-channel HFET phototransistor Q2 are coupled togetherand to the p-channel injector terminal of the thyristor 807.

A p-channel HFET phototransistor Q3 and an n-channel HFETphototransistor Q4 are coupled in series between the positive voltagesupply (V_(cc)) and the negative voltage supply (ground). The p-channelHFET phototransistor Q3 has a source and gate terminal both connected tothe positive voltage supply (V_(cc)). The n-channel HFET phototransistorQ4 has a source and gate terminal both connected to the negative voltagesupply (ground). The drain of the p-channel HFET phototransistor Q3 andthe drain of the n-channel HFET phototransistor Q4 are coupled togetherand to the n-channel injector terminal of the thyristor 807.

The digital optical signal A′ output from the Out0 output waveguide ofthe 180-degree optical hybrid coupler 800A is guided into the waveguideregion of the p-channel HFET phototransistor Q1 and the digital opticalsignal A′ output from the Out1 output waveguide of the 180-degreeoptical hybrid coupler 800A is guided into the waveguide region of thep-channel HFET phototransistor Q3. The digital optical signal B′ outputfrom the Out0 output waveguide of the 180-degree optical hybrid coupler800B is guided into the waveguide region of the n-channel HFETphototransistor Q2 and the digital optical signal B′ output from theOut1 output waveguide of the 180-degree optical hybrid coupler 800B isguided into the waveguide region of the n-channel HFET phototransistorQ4.

In this configuration, the p-channel HFET phototransistors Q1 and Q3 areactivated when the state of the digital optical signal A′ is ON, and then-channel HFET phototransistors Q2 and Q4 are activated when the stateof the digital optical signal B′ is ON. The p-channel HFETphototransistors Q1 is configured to operate as a turn-onphototransistor for the thyristor 807 when the state of the digitaloptical signal A′ is ON and the state of the digital optical signal B′is OFF. The n-channel HFET phototransistor Q4 is configured to operateas a turn-on phototransistor for the thyristor 807 when the state of thedigital optical signal B′ is ON and the state of the digital opticalsignal A′ is OFF. The p-channel HFET phototransistor Q3 and then-channel HFET phototransistor Q2 are configured to operate as turn-offphototransistors for the thyristor 807 when the states of both thedigital optical signal A′ and the digital optical signal B′ are ON.

Specifically, the phototransistors Q1, Q2, Q3 and Q4 can be sized sothat the turn-off phototransistors Q2 and Q3 are larger (for example, bya factor of two) than the turn-on phototransistors Q1 and Q4. When thestate of the digital optical signal A′ is ON and the state of thedigital optical signal B′ is OFF, the p-channel HFET phototransistors Q1and Q3 are activated and the n-channel HFET phototransistors Q2 and Q4are not activated. In this case, the turn-on p-channel HFETphototransistor Q1 supplies hole current to the p-channel injector andthe turn-off transistor Q3 draws electron current from the n-channelinjector; however, due to the internal gain of the thyristor, theinternal current flow of electrons is dominant which operates thethyristor 807 in its ON state where it conducts current verticallybetween the anode and the cathode terminal of the thyristor 807. Whenthe state of the digital optical signal A′ is OFF and the state of thedigital optical signal B′ is ON, the p-channel HFET phototransistors Q1and Q3 are not activated and the n-channel HFET phototransistors Q2 andQ4 are activated. In this case, the turn-on n-channel HFETphototransistor Q4 supplies electron current to the n-channel injectorand the turn-off transistor Q2 draws hole current from the p-channelinjector; however, due to the internal gain of the thyristor, theinternal current flow of holes is dominant which operates the thyristor807 in its ON state where it conducts current vertically between theanode and the cathode terminal of the thyristor 807. When the states ofboth the digital optical signal A′ and the digital optical signal B′ areON, the turn-on phototransistors Q1, Q2, Q3 and Q4 are all activated.However, in this case, the hole current drain operation of the largerturn-off transistor Q2 dominates the hole injection operation of thesmaller turn-on transistor Q1, which effectively operates to drain holecurrent from the p-channel injector of the thyristor 807. Similarly, theelectron current drain operation of the larger turn-off transistor Q3dominates the electron injection operation of the smaller turn-ontransistor Q4, which effectively operates to drain electron current fromthe n-channel injector of the thyristor 807. The drain of the holecurrent from the p-channel injector in conjunction with the drain ofelectron current from the n-channel injector operates the thyristor 807in its OFF state where there is minimal current conduction verticallybetween the anode and the cathode terminal of the thyristor 807. Whenthe states of both the digital optical signals A′ and B′ are OFF, noneof the phototransistors Q1, Q2, Q3, Q4 are activated and thyristor 807operates in its OFF state.

The ON and OFF states of the thyristor 807 generates the digitalelectrical signal C at the cathode terminal of the thyristor 808. The ONstate of the thyristor 807 can also be configured above the lasingthreshold such that the thyristor 807 emits a digital (ON/OFF) opticalsignal corresponding to the digital electrical signal C generated at thecathode terminal of the thyristor 807. In this configuration, the binarylevels of the digital electrical signal C (and possibly the ON/OFF stateof the corresponding digital optical signal emitted from the thyristor)for the ON and OFF states of the thyristor 807 corresponds to theBoolean XOR function of the ON/OFF states of the digital optical signalsA′ and B′ as shown in the table of FIG. 8B.

Optical 16-PSK Detector

An embodiment of the optical mixer and optical PSK detector 124A′″ fordetecting and demodulating optical 16-PSK signals is shown in FIGS. 9Aand 9B. It employs a network of six 180-degree optical hybrid couplers900A, 900B, 900C, 900D, 900E and 900F and four optical QPSK detectors124A′-1, 124A′-2, 124A′-3, 124A′-4 as described above with respect toFIG. 6A for optical mixing of the received phase-shifted carrier signaland the optical local oscillating signal output by the OPPL 122 anddemodulation.

As shown in FIG. 9A, waveguide 901A supplies the received phased-shiftedoptical carrier signal to the In0 input waveguide of the 180-degreeoptical hybrid coupler 900A. In this configuration, the 180-degreeoptical hybrid coupler 900A produces two output signals that propagateout from the Out0 and Out1 waveguides of the 180-degree optical hybridcoupler 900A. The intensity of the received phased-shifted opticalcarrier signal is split evenly (50:50 split) in each one of two outputsignals with a phase delay (lag) of 90 degrees (π/2 radians) for theoutput signal that propagates out from the Out1 waveguide relative theoutput signal that propagates out from the Out0 waveguide.

Waveguide 901B supplies the optical LO signal to the In0 input waveguideof the 180-degree optical hybrid coupler 900D. In this configuration,the 180-degree optical hybrid coupler 900D produces two output signalsthat propagate out from the Out0 and Out1 waveguides of the 180-degreeoptical hybrid coupler 900D. The intensity of the optical LO signal issplit evenly (50:50 split) in each one of two output signals with aphase delay (lag) of 90 degrees (π/2 radians) for the output signal thatpropagates out from the Out1 waveguide relative the output signal thatpropagates out from the Out0 waveguide.

Waveguide 903A supplies the optical signal output from the Out0 outputwaveguide of the 180-degree hybrid coupler 900A to an optical phasemodulator 905A, which functions to delay the phase of the suppliedreceived phased-shifted optical carrier signal by 67.5 degrees (or 3π/8radians). The resultant optical signal is supplied by waveguide 903C tothe In0 input waveguide of the 180-degree optical hybrid coupler 900B.In this configuration, the 180-degree optical hybrid coupler 900Bproduces two output signals that propagate out from the Out0 and Out1waveguides of the 180-degree optical hybrid coupler 900B. The intensityof the resultant optical signal is split evenly (50:50 split) in eachone of two output signals with a phase delay (lag) of 90 degrees (π/2radians) for the output signal that propagates out from the Out1waveguide relative the output signal that propagates out from the Out0waveguide. Waveguide 903D supplies the optical signal output from theOut0 output waveguide of the 180-degree hybrid coupler 900B to anoptical phase modulator 905B, which functions to delay the phase of thesupplied received phased-shifted optical carrier signal by 45 degrees(or π/4 radians). The resultant optical signal is supplied by waveguide903E to the input waveguide 601A of the optical QPSK detector 124A′-1.Waveguide 903F supplies the optical signal output from the Out1 outputwaveguide of the 180-degree hybrid coupler 900B to the input waveguide601A of the optical QPSK detector 124A′-2.

Waveguide 903B supplies the optical signal output from the Out1 outputwaveguide of the 180-degree hybrid coupler 900A to the In0 inputwaveguide of the 180-degree optical hybrid coupler 900C. In thisconfiguration, the 180-degree optical hybrid coupler 900C produces twooutput signals that propagate out from the Out0 and Out1 waveguides ofthe 180-degree optical hybrid coupler 900C. The intensity of theresultant optical signal is split evenly (50:50 split) in each one oftwo output signals with a phase delay (lag) of 90 degrees (π/2 radians)for the output signal that propagates out from the Out1 waveguiderelative the output signal that propagates out from the Out0 waveguide.Waveguide 903G supplies the optical signal output from the Out0 outputwaveguide of the 180-degree hybrid coupler 900C to an optical phasemodulator 905C, which functions to delay the phase of the suppliedreceived phased-shifted optical carrier signal by 45 degrees (or π/4radians). The resultant optical signal is supplied by waveguide 903H tothe input waveguide 601A of the optical QPSK detector 124A′-3. Waveguide903I supplies the optical signal output from the Out1 output waveguideof the 180-degree hybrid coupler 900C to the input waveguide 601A of theoptical QPSK detector 124A′-4.

Waveguide 903J supplies the optical LO signal output from the Out0output waveguide of the 180-degree hybrid coupler 900D to the In0 inputwaveguide of the 180-degree optical hybrid coupler 900E. In thisconfiguration, the 180-degree optical hybrid coupler 900E produces twooutput signals that propagate out from the Out0 and Out1 waveguides ofthe 180-degree optical hybrid coupler 900E. The intensity of theresultant optical signal is split evenly (50:50 split) in each one oftwo output signals with a phase delay (lag) of 90 degrees (π/2 radians)for the output signal that propagates out from the Out1 waveguiderelative the output signal that propagates out from the Out0 waveguide.Waveguide 903L supplies the optical signal output from the Out0 outputwaveguide of the 180-degree hybrid coupler 900E to the input waveguide601B of the optical QPSK detector 124A′-1. Waveguide 903M supplies theoptical signal output from the Out1 output waveguide of the 180-degreehybrid coupler 900E to the input waveguide 601B of the optical QPSKdetector 124A′-2.

Waveguide 903K supplies the optical LO signal output from the Out1output waveguide of the 180-degree hybrid coupler 900D to the In0 inputwaveguide of the 180-degree optical hybrid coupler 900F. In thisconfiguration, the 180-degree optical hybrid coupler 900F produces twooutput signals that propagate out from the Out0 and Out1 waveguides ofthe 180-degree optical hybrid coupler 900F. The intensity of theresultant optical signal is split evenly (50:50 split) in each one oftwo output signals with a phase delay (lag) of 90 degrees (π/2 radians)for the output signal that propagates out from the Out1 waveguiderelative the output signal that propagates out from the Out0 waveguide.Waveguide 903N supplies the optical signal output from the Out0 outputwaveguide of the 180-degree hybrid coupler 900F to the input waveguide601B of the optical QPSK detector 124A′-3. Waveguide 903O supplies theoptical signal output from the Out1 output waveguide of the 180-degreehybrid coupler 000F to the input waveguide 601B of the optical QPSKdetector 124A′-4.

In this configuration, the phase of the received phase-shifted opticalcarrier signal that is supplied via the waveguide 903E to the inputwaveguide 601A of the optical QPSK detector 124A′-1 lags by 112.5degrees (5π/8 radians) the phase of the received phase-shifted opticalcarrier signal. The phase of the received phase-shifted optical carriersignal that is supplied via the waveguide 903F to the input waveguide601A of the optical QPSK detector 124A′-2 lags by 157.5 degrees (7π/8radians) the phase of the received phase-shifted optical carrier signal.The phase of the received phase-shifted optical carrier signal that issupplied via the waveguide 903H to the input waveguide 601A of theoptical QPSK detector 124A′-3 lags by 135 degrees (3π/4 radians) thephase of the received phase-shifted optical carrier signal. The phase ofthe received phase-shifted optical carrier signal that is supplied viathe waveguide 903I to the input waveguide 601A of the optical QPSKdetector 124A′-4 lags by 180 degrees (π radians) the phase of thereceived phase-shifted optical carrier signal. The phase of the opticalLO signal that is supplied via the waveguide 903L to the input waveguide601B of optical QPSK detector 124A′-1 is aligned (lag near zero degreesor radians) to the phase of the optical LO signal supplied by the OPLL122. The phase of the optical LO signal that is supplied via thewaveguide 903M to the input waveguide 601B of optical QPSK detector124A′-2 lags by 90 degrees (π/2 radians) the phase of the optical LOsignal that is supplied by the OPLL 122. The phase of the optical LOsignal that is supplied via the waveguide 903N to the input waveguide601B of optical QPSK detector 124A′-3 lags by 90 degrees (π/2 radians)the phase of the optical LO signal that is supplied by the OPLL 122. Thephase of the optical LO signal that is supplied via the waveguide 903Oto the input waveguide 601B of optical QPSK detector 124A′-4 lags by 180degrees (π radians) the phase of the optical LO signal that is suppliedby the OPLL 122.

Furthermore, in this configuration, the ON and OFF states of thethyristors of the optical QPSK detector 124A′-1 generates digitalelectrical signals A, Ā, B, B (and possibly corresponding digitaloptical signals A, B). The ON and OFF states of the thyristors of theoptical QPSK detector 124A′-2 generates digital electrical signals A′,Ā′, B′, B′ (and possibly corresponding digital optical signals A′, B′).The ON and OFF states of the thyristors of the optical QPSK detector124A′-3 generates digital electrical signals a, ā, b, b (and possiblycorresponding digital optical signals a, b). The ON and OFF states ofthe thyristors of the optical QPSK detector 124A′-4 generates digitalelectrical signals a′, ā′, b′, b′ (and possibly corresponding digitaloptical signals a′, b′).

As shown in FIG. 9B, the digital optical signals A′, B′ are supplied toa thyristor-based optical XOR block 907A that generates a digitalelectrical signal C (and possibly a corresponding digital optical signalC) that is the Boolean XOR function of the ON/OFF states of the digitaloptical signals A′, B′. The digital optical signals a, b are supplied toa thyristor-based optical XOR block 907B that generates a digitaloptical signal c (and possibly a corresponding digital optical signal c)that is the Boolean XOR function of the ON/OFF states of the digitaloptical signals a, b. The digital optical signals a′, b′ are supplied toa thyristor-based optical XOR block 907C that generates a digitaloptical signal c′ (and possibly a corresponding digital optical signalc) that is the Boolean XOR function of the ON/OFF states of the digitaloptical signals a′, b′. The digital optical signals c, c′ are suppliedto a thyristor-based optical XOR block 907D that generates a digitalelectrical signal D (and possibly a corresponding digital optical signalD) that is the Boolean XOR function of the ON/OFF states of the digitaloptical signals c, c′. The optical XOR blocks 907A, 907B, 907D and 907Dcan be realized with the optical XOR block described above with respectto FIGS. 8A and 8B.

The binary levels of the digital electrical signals A, B, C, Dcorrespond to the sixteen possible phase offsets of the receivedphase-shifted optical carrier signal. An example of correspondencebetween the binary levels of the digital electrical signals A, B, C, Dand four phase offsets of the received phase-shifted optical carriersignal is shown in the chart of FIG. 9C and the phase constellationdiagram of FIG. 9D. Note that these sixteen possible phase offsets are22.5 degrees (or π/8 radians) apart from one another.

Optical 16-QAM Detector

An embodiment of the optical mixer and optical PSK detector 124A″″ fordetecting and demodulating optical 16-QAM signals is shown in FIGS. 10Aand 10B. It employs a network of six 180-degree optical hybrid couplers1000A, 1000B, 1000C, 1000D, 1000E and 1000F and four optical QPSKdetectors 124A′-1, 124A′-2, 124A′-3, 124A′-4 as described above withrespect to FIG. 6A for optical mixing of the received phase-shiftedcarrier signal and the optical local oscillating signal output by theOPPL 122 and demodulation. The optical 16-QAM signals employ acombination of amplitude modulation (with four possible levels ofamplitude modulation increasing in magnitude and designated T1, T2, T3and T4) and PSK modulation (with four possible phase offsets peramplitude modulation level). The Q1 and Q4 phototransistors of the fouroptical QPSK detectors operate to inject hole/electron current intocorresponding thyristor injector terminals. Similarly, the Q2 and Q3phototransistors of the four optical QPSK detectors operate to drainhole/electron current from the corresponding thyristor injectorterminals. Furthermore, the supply voltages supplied to the thyristorsof the four optical QPSK detectors are adjusted such that thresholdlevels for activating such thyristors increases in magnitude over thefour optical QPSK detectors to correspond to the four possible levels ofamplitude modulation T1, T2, T3 and T4. In this manner, the thresholdlevel for activation of the thyristor for the optical QPSK detector124A′-1 is less than the threshold level for activation of the thyristorfor the optical QPSK detector 124A′-2, which is less than the thresholdlevel for activation of the thyristor for the optical QPSK detector124A′-3, which is less than the threshold level for activation of thethyristor for the optical QPSK detector 124A′-4. This configurationprovides for optical detection and demodulation of the optical 16-QAMsignals.

As shown in FIG. 10A, waveguide 1001A supplies the receivedphased-shifted optical carrier signal to the In0 input waveguide of the180-degree optical hybrid coupler 1000A. In this configuration, the180-degree optical hybrid coupler 1000A produces two output signals thatpropagate out from the Out0 and Out1 waveguides of the 180-degreeoptical hybrid coupler 1000A. The intensity of the receivedphased-shifted optical carrier signal is split evenly (50:50 split) ineach one of two output signals with a phase delay (lag) of 90 degrees(π/2 radians) for the output signal that propagates out from the Out1waveguide relative the output signal that propagates out from the Out0waveguide.

Waveguide 1001B supplies the optical LO signal to the In0 inputwaveguide of the 180-degree optical hybrid coupler 1000D. In thisconfiguration, the 180-degree optical hybrid coupler 1000D produces twooutput signals that propagate out from the Out0 and Out1 waveguides ofthe 180-degree optical hybrid coupler 1000D. The intensity of theoptical LO signal is split evenly (50:50 split) in each one of twooutput signals with a phase delay (lag) of 90 degrees (π/2 radians) forthe output signal that propagates out from the Out1 waveguide relativethe output signal that propagates out from the Out0 waveguide.

Waveguide 1003A supplies the optical signal output from the Out0 outputwaveguide of the 180-degree hybrid coupler 1000A to the In0 inputwaveguide of the 180-degree optical hybrid coupler 1000B. In thisconfiguration, the 180-degree optical hybrid coupler 1000B produces twooutput signals that propagate out from the Out0 and Out1 waveguides ofthe 180-degree optical hybrid coupler 1000B. The intensity of theresultant optical signal is split evenly (50:50 split) in each one oftwo output signals with a phase delay (lag) of 90 degrees (π/2 radians)for the output signal that propagates out from the Out1 waveguiderelative the output signal that propagates out from the Out0 waveguide.Waveguide 1003C supplies the optical signal output from the Out0 outputwaveguide of the 180-degree hybrid coupler 1000B to the input waveguide601A of the optical QPSK detector 124A′-1. Waveguide 1003D supplies theoptical signal output from the Out1 output waveguide of the 180-degreehybrid coupler 1000B to the input waveguide 601A of the optical QPSKdetector 124A′-2.

Waveguide 1003B supplies the optical signal output from the Out1 outputwaveguide of the 180-degree hybrid coupler 1000A to the In0 inputwaveguide of the 180-degree optical hybrid coupler 1000C. In thisconfiguration, the 180-degree optical hybrid coupler 1000C produces twooutput signals that propagate out from the Out0 and Out1 waveguides ofthe 180-degree optical hybrid coupler 1000C. The intensity of theresultant optical signal is split evenly (50:50 split) in each one oftwo output signals with a phase delay (lag) of 90 degrees (π/2 radians)for the output signal that propagates out from the Out1 waveguiderelative the output signal that propagates out from the Out0 waveguide.Waveguide 1003E supplies the optical signal output from the Out0 outputwaveguide of the 180-degree hybrid coupler 1000C to the input waveguide601A of the optical QPSK detector 124A′-3. Waveguide 1003F supplies theoptical signal output from the Out1 output waveguide of the 180-degreehybrid coupler 1000C to the input waveguide 601A of the optical QPSKdetector 124A′-4.

Waveguide 1003G supplies the optical LO signal output from the Out0output waveguide of the 180-degree hybrid coupler 1000D to the In0 inputwaveguide of the 180-degree optical hybrid coupler 1000E. In thisconfiguration, the 180-degree optical hybrid coupler 1000E produces twooutput signals that propagate out from the Out0 and Out1 waveguides ofthe 180-degree optical hybrid coupler 1000E. The intensity of theresultant optical signal is split evenly (50:50 split) in each one oftwo output signals with a phase delay (lag) of 90 degrees (π/2 radians)for the output signal that propagates out from the Out1 waveguiderelative the output signal that propagates out from the Out0 waveguide.Waveguide 1003I supplies the optical signal output from the Out0 outputwaveguide of the 180-degree hybrid coupler 1000E to the input waveguide601B of the optical QPSK detector 124A′-1. Waveguide 1003J supplies theoptical signal output from the Out1 output waveguide of the 180-degreehybrid coupler 1000E to the input waveguide 601B of the optical QPSKdetector 124A′-2.

Waveguide 1003H supplies the optical LO signal output from the Out1output waveguide of the 180-degree hybrid coupler 1000D to the In0 inputwaveguide of the 180-degree optical hybrid coupler 1000F. In thisconfiguration, the 180-degree optical hybrid coupler 1000F produces twooutput signals that propagate out from the Out0 and Out1 waveguides ofthe 180-degree optical hybrid coupler 1000F. The intensity of theresultant optical signal is split evenly (50:50 split) in each one oftwo output signals with a phase delay (lag) of 90 degrees (π/2 radians)for the output signal that propagates out from the Out1 waveguiderelative the output signal that propagates out from the Out0 waveguide.Waveguide 1003K supplies the optical signal output from the Out0 outputwaveguide of the 180-degree hybrid coupler 1000F to the input waveguide601B of the optical QPSK detector 124A′-3. Waveguide 1003L supplies theoptical signal output from the Out1 output waveguide of the 180-degreehybrid coupler 1000F to the input waveguide 601B of the optical QPSKdetector 124A′-4.

In this configuration, the phase of the received phase-shifted opticalcarrier signal that is supplied via the waveguide 1003C to the inputwaveguide 601A of the optical QPSK detector 124A′-1 is aligned (lag nearzero degrees or radians) to the phase of the received phase-shiftedoptical carrier signal. The phase of the received phase-shifted opticalcarrier signal that is supplied via the waveguide 1003D to the inputwaveguide 601A of the optical QPSK detector 124A′-2 lags by 90 degrees(π/2 radians) the phase of the received phase-shifted optical carriersignal. The phase of the received phase-shifted optical carrier signalthat is supplied via the waveguide 1003E to the input waveguide 601A ofthe optical QPSK detector 124A′-3 lags by 90 degrees (π/2 radians) thephase of the received phase-shifted optical carrier signal. The phase ofthe received phase-shifted optical carrier signal that is supplied viathe waveguide 1003F to the input waveguide 601A of the optical QPSKdetector 124A′-4 lags by 180 degrees (π radians) the phase of thereceived phase-shifted optical carrier signal. The phase of the opticalLO signal that is supplied via the waveguide 1003I to the inputwaveguide 601B of optical QPSK detector 124A′-1 is aligned (lag nearzero degrees or radians) to the phase of the optical LO signal suppliedby the OPLL 122. The phase of the optical LO signal that is supplied viathe waveguide 1003J to the input waveguide 601B of optical QPSK detector124A′-2 lags by 90 degrees (π/2 radians) the phase of the optical LOsignal that is supplied by the OPLL 122. The phase of the optical LOsignal that is supplied via the waveguide 1003K to the input waveguide601B of optical QPSK detector 124A′-3 lags by 90 degrees (π/2 radians)the phase of the optical LO signal that is supplied by the OPLL 122. Thephase of the optical LO signal that is supplied via the waveguide 1003Lto the input waveguide 601B of optical QPSK detector 124A′-4 lags by 180degrees (π radians) the phase of the optical LO signal that is suppliedby the OPLL 122.

As described above, the Q1 and Q4 phototransistors of the four opticalQPSK detectors operate to inject hole/electron current intocorresponding thyristor injector terminals. Similarly, the Q2 and Q3phototransistors of the four optical QPSK detectors operate to drainhole/electron current from the corresponding thyristor injectorterminals. Furthermore, the supply voltages supplied to the thyristorsof the four optical QPSK detectors are adjusted such that thresholdlevels for activating such thyristors increases in magnitude over thefour optical QPSK detectors to correspond to the four possible levels ofamplitude modulation T1, T2, T3 and T4.

In this configuration, the ON and OFF states of the thyristors of theoptical QPSK detector 124A′-1 generates digital electrical signals A1,A1 , B1, B1 (and possibly corresponding digital optical signals A1, B1)corresponding to four possible phase offsets for the T1 level ofamplitude modulation. The ON and OFF states of the thyristors of theoptical QPSK detector 124A′-2 generates digital electrical signals A2,A2 , B2, B2 (and possibly corresponding digital optical signals A2, B2)corresponding to four possible phase offsets for the T2 level ofamplitude modulation. The ON and OFF states of the thyristors of theoptical QPSK detector 124A′-3 generates digital electrical signals A3,A3 , B3, B3 (and possibly corresponding digital optical signals A3, B3)corresponding to four possible phase offsets for the T3 level ofamplitude modulation. The ON and OFF states of the thyristors of theoptical QPSK detector 124A′-4 generates digital electrical signals A4,A4 , B4, B4 (and possibly corresponding digital optical signals A4, B4)corresponding to four possible phase offsets for the T4 level ofamplitude modulation. In this manner, the binary levels of theelectrical signals A1, B1, A2, B2, A3, B3, A4, B4 correspond to thesixteen possible combinations of amplitude modulation levels(T1/T2/T3/T4) and phase offsets of the received optical carrier signal.

An example of correspondence between the binary levels of the digitalelectrical signals A1, B1, A2, B2, A3, B3, A4, B4 and sixteencombinations of amplitude modulation levels (T1/T2/T3/T4) and phaseoffsets of the received optical carrier signal is shown in the charts ofFIGS. 10C and 10E and the phase constellation diagrams of FIGS. 10D and10F. The chart of FIG. 10C and the constellation diagram of FIG. 10Dshow an example of correspondence between the binary levels of theelectrical signals A1, B1, A2, B2 and eight combinations of amplitudemodulation levels (T1/T2) and phase offsets of the received opticalcarrier signal.

FIG. 10E shows an embodiment for one part (the BPSK detector for the A2signal) of the optical QPSK detector 124A′-2 for the T2 level ofamplitude modulation. In this embodiment, the voltage supply potential(V_(T2)) is configured to set the switching threshold of the thyristor307 (the point where the thyristor 307 operates in its ON state where itconducts current vertically between the anode and the cathode terminalof the thyristor 307) such that is larger than the switching thresholdfor the thyristor 307 of the optical QPSK detector 124A′-1 for the T1level of amplitude modulation. A similar configuration is used for theother part (the BPSK detector for the B2 signal) of the optical QPSKdetector 124A′-2 for the T2 level of amplitude modulation.

The chart of FIG. 10F and the constellation diagram of FIG. 10G show anexample of correspondence between the binary levels of the digitalelectrical signals A3, B3, A4, B4 and eight combinations of amplitudemodulation levels (T3/T4) and phase offsets of the received opticalcarrier signal.

FIG. 10H shows an embodiment for one part (the BPSK detector for the A3signal) of the optical QPSK detector 124A′-3 for the T3 level ofamplitude modulation. In this embodiment, the voltage supply potential(V_(T3)) is configured to set the switching threshold of the thyristor307 such that is larger than the switching threshold for the thyristor307 of the optical QPSK detector 124A′-2 for the T2 level of amplitudemodulation. A similar configuration is used for the other part (the BPSKdetector for the B3 signal) of the optical QPSK detector 124A′-3 for theT3 level of amplitude modulation.

FIG. 10I shows an embodiment for one part (the BPSK detector for the A4signal) of the optical QPSK detector 124A′-4 for the T4 level ofamplitude modulation. In this embodiment, the voltage supply potential(V_(T4)) is configured to set the switching threshold of the thyristor307 such that is larger than the switching threshold for the thyristor307 of the optical QPSK detector 124A′-2 for the T3 level of amplitudemodulation. A similar configuration is used for the other part (the BPSKdetector for the B4 signal) of the optical QPSK detector 124A′-4 for theT4 level of amplitude modulation.

FIG. 10J shows the combination of the constellation diagrams of FIGS.10D and 10G to show the sixteen possible combinations of amplitudemodulation levels (T1/T2/T3/T4) and phase offsets of the receivedoptical carrier signal that can be represented by the binary levels ofthe digital electrical signals A1, B1, A2, B2, A3, B3, A4, B4. Note thatthe four possible phase offsets for each amplitude modulation level are90 degrees (or π/2 radians) apart from one another.

Optical Phase Lock Loop/Clock Recovery Block

FIG. 11 is a block diagram illustrating an embodiment of the clockrecovery block 124 of FIG. 2B (and FIG. 5), which is realized by anoptical phase lock loop that includes a phase-tunable optoelectronicoscillator 1101, an optical phase detector 1103 and an optical chargepump and loop filter circuit 1105. The clock recovery block(particularly the optical phase detector 1103) is supplied with areference optical signal input (in this example, the digital opticalsignal input 1111 generated by the PSK detector, which represents arecovered signal stream of symbols carried by a received phase-shiftedoptical carrier signal). The symbol clock employed by a transmitter inmodulating the phase of the optical carrier signal is embedded in thisdigital optical signal input 1111. The phase-tunable optoelectronicoscillator 1101 of the clock recovery block is configured to produce anoptical clock signal (and corresponding electrical clock signal) havingON/OFF transitions that are synchronized to the timing of the referenceoptical signal input (in this example, the timing of symbol clock of thetransmitter embedded in the digital optical signal input 1111).

The optoelectronic oscillator 1101 produces an optical clock signal1109A and corresponding electrical clock signal 1109B upon activation bya start signal 1107. The phase of both the optical clock signal 1109Aand the corresponding electrical clock signal 1109B is tuned accordingto one or more phase tuning signals 1115 that are supplied to theoptoelectronic oscillator 1101. The digital optical signal input 1111generated by the PSK detector as well as the optical clock signal 1109Aproduced by the optoelectronic oscillator 1101 are supplied bywaveguides to the optical phase detector 1103, which functions todetermine the digital phase difference (or equivalently to determine thedifference in digital phase) between the digital optical signal input1111 and the optical clock signal 1109A and outputs one or more opticalphase difference signals 1113 representative of such phase difference tothe optical charge pump and loop filter 1105. The optical charge pumpand loop filter circuit 1005 functions as a loop filter to integrate theoptical phase difference signals 1113 over time and generate the phasetuning signal(s) 1115 for control of the phase of the optical clocksignal 1109A (as well as the phase of the corresponding electrical clocksignal 1109B) such that it matches the phase of the symbol clock signalembedded within the digital optical signal input 1111. In the lockedstate where the phase of the optical clock signal 1109A (and the phaseof the corresponding electrical clock signal 1109B) matches the phase ofthe symbol clock signal, the frequency of the optical clock signal 1109A(and the frequency of the corresponding electrical clock signal 1109B)matches the frequency of the symbol clock signal. In this manner, theclock recovery block 124 operates to automatically tune the frequencyand phase of the optical clock signal 1109A (and the frequency and phaseof the corresponding electrical clock signal 1109B) into this lockedstate in order to recover the symbol clock signal embedded in thedigital optical signal input 1111.

The optical phase lock loop of FIG. 11 can be adapted to perform a widevariety of other functions beyond clock recovery. Such applications caninclude clock generation (such as up-conversion of a low frequency clockto a higher frequency clock or down-conversion of a higher frequencyclock to a lower frequency clock), generation of local oscillatingsignals for communication systems (such as generation of localoscillating signals for modulation and demodulation), and clockdistribution and jitter compensation in integrated circuits,instrumentation systems and communication systems.

Optical Phase Detector

FIG. 12A illustrates an embodiment of the optical phase detector 1103 ofFIG. 11, which includes an AND gate 1205 operably coupled between twooptical flip-flops 1203A and 1203B. The optical flip-flop 1203A receivesthe digital optical signal input (D_(A)) 1111 at its D_(A) input andfunctions to generate a digital optical signal output Q_(A) andcorresponding digital electrical signal output Q_(A) based on the binarylevel of the complementary digital electrical Reset/Reset signals at itsR/R inputs and the digital optical signal input (D_(A)) 1111 at itsD_(A) input according to the table of FIG. 12B. The optical flip-flop1203B receives the optical clock signal (D_(B)) 1109A at its D_(B) inputand functions to generate a digital optical signal output Q_(B) andcorresponding digital electrical signal output Q_(B) based on the binarylevel of the complementary digital electrical Reset/Reset signals at itsR/R inputs and the clock signal (D_(B)) 1109A at its D_(B) inputaccording to table of FIG. 12C. The AND gate 1205 receives the digitalelectrical output signals Q_(A) and Q_(B) output by the opticalflip-flops 1203A and 1203B and generates the complementary electricalReset/Reset signals according to the table of FIG. 12D. Thecomplementary electrical Reset/Reset signals are supplied to the R and Rinputs of both optical flip-flops 1203A and 1203B.

In this configuration, the ON level of the digital optical signal inputQ_(A) (and the corresponding high level of the digital electrical signaloutput Q_(A)) represents the digital phase difference between thedigital optical signal input 1111 and the optical clock signal 1109A(where the digital optical signal input 1111 is ON and the optical clocksignal 1109A is OFF). Similarly, the ON level of the digital opticalsignal output Q_(B) (and the corresponding high level of the electricalsignal Q_(B)) represented the digital phase difference between theoptical clock signal 1109A and the digital optical signal input 1111(where the optical clock signal 1109A is ON and the digital opticalsignal input 1111 is OFF). The digital electrical and optical signalQ_(A) as well as the digital electrical and optical signals Q_(B) areoutput to the optical charge pump and filter circuit 1105.

The optical charge pump and filter circuit 1105 operates as a loopfilter that converts the digital optical signal Q_(A) into acorresponding voltage signal that tracks whether the digital opticalsignal output Q_(A) (and the corresponding digital electrical signalQ_(A)) is greater than the digital optical signal output Q_(B) (and thecorresponding digital electrical signal Q_(B)) over time. This output isused as a phase tuning signal 1115 supplied to the optoelectronicoscillator 1101 in order to increase the frequency of the optical clocksignal 1109A and the corresponding electrical clock signal 1109Bgenerated by the optoelectronic oscillator 1101. The optical charge pumpand loop filter circuit 1105 also operates as a loop filter thatconverts the digital optical signal Q_(B) into a corresponding voltagesignal that tracks whether the digital optical signal output Q_(B) (andthe corresponding digital electrical signal Q_(B)) is greater than thedigital optical signal output Q_(A) (and the corresponding digitalelectrical signal Q_(A)) over time. This output is used as a phasetuning signal 1115 supplied to the optoelectronic oscillator 1101 inorder to decrease the frequency of the optical clock signal 1109A andthe corresponding electrical clock signal 1109B generated by theoptoelectronic oscillator 1101. In this configuration, the frequency andphase of the optical clock signal 1109A and the corresponding electricalclock signal 1109B generated by the optoelectronic oscillator 1101 willbe adjusted such that it matches the frequency and phase of the symbolclock for the recovered digital optical signal input 1111.

FIG. 13A illustrates an embodiment of the optical flip-flop 1203A ofFIG. 12A. The optical flip-flop is a circuit that has two stable statesand can be used to store state information. The circuit can be made tochange state by signals applied to one or more control inputs (where atleast one control input is an optical signal) and will have one or moreoutputs (where one or more of such outputs can be an optical signal). Itcan be used as basic storage element in sequential logic. It also can beused as a fundamental building block of digital electronics systems usedin computers, communications, and many other types of systems. When usedin a finite-state machine, the output and next state depend not only onits current input, but also on its current state (and hence, previousinputs). It can also be used for counting of pulses, and forsynchronizing variably-timed input signals to some reference timingsignal.

A waveguide (not shown), which is the D_(A) input of the opticalflip-flop 1203A, supplies an optical control input signal (in thisexample, the recovered digital optical signal input 1111) to thewaveguide region of a four terminal vertical thyristor (N region-Pregion-N region-P region) 1300 with a split load. A first load element1301A is coupled between the positive voltage supply (V_(cc)) and theanode terminal of the thyristor 1300. A second load element 1301B iscoupled between the cathode terminal of the thyristor 1300 and thenegative voltage supply (ground). The thyristor 1300 also has ap-channel injector terminal and an n-channel injector terminal. Ap-channel HFET transistor Q1 is coupled between the positive voltagesupply (V_(cc)) and the n-channel injector terminal. An n-channel HFETtransistor Q2 is coupled between the negative voltage supply (ground)and the p-channel injector terminal. Two complementary electricalcontrol inputs (Reset/Reset signals), which in this example are outputby the AND gate 1205, are supplied as inputs to the optical flip-flop1203A. The electrical Reset signal is supplied as an input to the gateterminal of the p-channel HFET transistor Q1. The electrical Resetsignal is supplied as an input to the gate terminal of the n-channelHFET transistor Q2. The p-channel HFET transistor Q1 is activated andoperates as a turn-off transistor for the thyristor 1300 when the levelof the electrical Reset signal is low. The n-channel HFET transistor Q2is activated and operates as a turn-off transistor for the thyristor1300 when the level of the electrical Reset signal is high. Thecomplementary nature of the electrical Reset and Reset signals dictatesthat the p-channel HFET transistor Q1 and the n-channel HFET transistorQ2 are activated at the same time when the level of the electrical Resetsignal is low and the level of the complementary electrical Reset signalis high. When activated, the p-channel HFET transistor Q1 drainselectron current from (i.e., supplies hole current to) the n-channelinjector and the n-channel HFET transistor Q2 drains hole current from(i.e., supplies electron current to) the p-channel injector of thethyristor 1300 such that the thyristor 1300 switches OFF and there isminimal current conduction between the anode terminal and the cathodeterminal of the thyristor 1300. When the electrical Reset signal is high(and the complementary electrical Reset signal is low) and the opticalcontrol input signal (in this example, the recovered digital opticalsignal input 1111) is ON, the optical control input signal is absorbedby the thyristor 1300 in order to introduce majority charge carriersinto the thyristor 1300. Such absorbed charge is sufficient to operatethe thyristor 1300 in its ON state where it conducts current verticallybetween the anode terminal and the cathode terminal of the thyristor1300. In this ON state, the current that conducts vertically between theanode terminal and the cathode terminal is above the lasing threshold ofthe thyristor 1300 in order to produce a digital optical signal Q_(A)that is emitted from the thyristor 1300. A corresponding digitalelectrical signal Q_(A) is produced at the cathode terminal of thethyristor 1300. Note that the current draining operations of the HFETtransistors Q1 and Q2 that operate the thyristor 1300 in its OFF statein response to the electrical Reset signal being low (and thecomplementary electrical Reset signal being high) overrides the ON stateoperation of the thyristor 1300 triggered by the ON state of the opticalcontrol input signal. This arises because the HFET transistors Q1 and Q2force the OFF state of the thyristor 1300. In this manner, the ON andOFF states of the thyristor 1300 emits a digital (ON/OFF) optical signalQ_(A) and generates a corresponding digital electrical signal Q_(A) atthe cathode terminal of the thyristor 1300. The ON/OFF states of thedigital optical signal Q_(A) and corresponding binary levels of thedigital electrical signal Q_(A) correspond to the binary levels of therecovered digital optical signal 1111 (D_(A)) and the complementaryelectrical Reset/Reset signals according to the table of FIG. 12B.

FIG. 13B illustrates an embodiment of the optical flip-flop 1203B ofFIG. 12A similar to the optical flip-flip of FIG. 13A. A waveguide (notshown), which is the D_(B) input of the optical flip-flop 1203B,supplies an optical control input signal (in this example, the opticalclock signal 1009A) to the waveguide region of a four terminal verticalthyristor (N region-P region-N region-P region) 1302 with a split load.A first load element 1303A is coupled between the positive voltagesupply (V_(cc)) and the anode terminal of the thyristor 1302. A secondload element 1303B is coupled between the cathode terminal of thethyristor 1303 and the negative voltage supply (ground). The thyristor1302 also has a p-channel injector terminal and an n-channel injectorterminal. A p-channel HFET transistor Q1 is coupled between the positivevoltage supply (V_(cc)) and the n-channel injector terminal. Ann-channel HFET transistor Q2 is coupled between the negative voltagesupply (ground) and the p-channel injector terminal. Two complementaryelectrical control inputs (Reset/Reset signals), which in this exampleare output by the AND gate 1205, are supplied as inputs to the opticalflip-flop 1203B. The electrical Reset signal is supplied as an input tothe gate terminal of the p-channel HFET transistor Q1. The electricalReset signal is supplied as an input to the gate terminal of then-channel HFET transistor Q2. The p-channel HFET transistor Q1 isactivated and operates as a turn-off transistor for the thyristor 1302when the level of the electrical Reset signal is low. The n-channel HFETtransistor Q2 is activated and operates as a turn-off transistor for thethyristor 1302 when the level of the electrical Reset signal is high.The complementary nature of the electrical Reset and Reset signalsdictates that the p-channel HFET transistor Q1 and the n-channel HFETtransistor Q2 are activated at the same time when the level of theelectrical Reset signal is low and the level of the complementaryelectrical Reset signal is high. When activated, the p-channel HFETtransistor Q1 drains electron current from (i.e., supplies hole currentto) the n-channel injector and the n-channel HFET transistor Q2 drainshole current from (i.e., supplies electron current to) the p-channelinjector of the thyristor 1302 such that the thyristor 1302 switches OFFand there is minimal current conduction between the anode terminal andthe cathode terminal of the thyristor 1302. When the electrical Resetsignal is high (and the complementary electrical Reset signal is low)and the optical control input signal (in this example, optical clocksignal 1009A) is ON, the optical control input signal is absorbed by thethyristor 1302 in order to introduce majority charge carriers into thethyristor 1302. Such absorbed charge is sufficient to operate thethyristor 1302 in its ON state where it conducts current verticallybetween the anode terminal and the cathode terminal of the thyristor1302. In this ON state, the current that conducts vertically between theanode terminal and the cathode terminal is above the lasing threshold ofthe thyristor 1302 in order to produce a digital optical signal Q_(B)that is emitted from the thyristor 1302. A corresponding digitalelectrical signal Q_(B) is produced at the cathode terminal of thethyristor 1302. Note that the current draining operations of the HFETtransistors Q1 and Q2 that operate the thyristor 1302 in its OFF statein response to the electrical Reset signal being low (and thecomplementary electrical Reset signal being high) overrides the ON stateoperation of the thyristor 1302 triggered by the ON state of the opticalcontrol input signal. This arises because the HFET transistors Q1 and Q2force the OFF state of the thyristor 1302. In this manner, the ON andOFF states of the thyristor 1302 emits a digital (ON/OFF) optical signalQ_(B) and generates a corresponding digital electrical signal Q_(B) atthe cathode terminal of the thyristor 1302. The ON/OFF states of thedigital optical signal Q_(B) and corresponding binary levels of thedigital electrical signal Q_(B) correspond to the ON/OFF states of theoptical clock signal 1009A (D_(B)) and the complementary electricalReset/Reset signals according to the table of FIG. 12C.

FIG. 14A illustrates an embodiment of an AND gate 1205 of FIG. 12 withelectrical inputs. The AND gate 1205 is basic digital logic gateproduces a digital electrical output signal (in this example, theelectrical Reset signal) that is the logical conjunction (the BooleanAND function) of two digital electrical inputs (in this example, thedigital electrical signals Q_(A) and Q_(B)) according to the table ofFIG. 12D. The AND gate 1205 includes a four terminal vertical thyristor(N region-P region-N region-P region) 1400 with a split load. A firstload element 1401A is coupled between the positive voltage supply(V_(cc)) and the anode terminal of the thyristor 1400. A second loadelement 1401B is coupled between the cathode terminal of the thyristor1400 and the negative voltage supply (ground). The thyristor 1400 alsohas a p-channel injector terminal and an n-channel injector terminal.

A p-channel HFET transistor Q1 and an n-channel HFET transistor Q2 arecoupled in series between the positive voltage supply (V_(cc)) and thenegative voltage supply (ground). The p-channel HFET transistor Q1 has asource terminal connected to the positive voltage supply (V_(cc)). Then-channel HFET transistor Q2 has a source terminal connected to thenegative voltage supply (ground). The drain terminal of the p-channelHFET transistor Q1 and the drain terminal of the n-channel HFETtransistor Q2 are coupled together and to the n-channel injectorterminal of the thyristor 1400. An n-channel HFET transistor Q3 and ap-channel HFET transistor Q4 are coupled in series between the positivevoltage supply (V_(cc)) and the negative voltage supply (ground). Then-channel HFET transistor Q3 has a drain terminal connected to thepositive voltage supply (V_(cc)). The p-channel HFET transistor Q4 has adrain terminal connected to the negative voltage supply (ground). Thesource terminal of the n-channel HFET transistor Q3 and the sourceterminal of the p-channel HFET transistor Q4 are coupled together and tothe p-channel injector terminal of the thyristor 1400.

The digital electrical input Q_(A) (which is supplied from the opticalflip-flop 1203A in this example) is supplied as an input to the gateterminals of both the p-channel HFET transistor Q1 and the n-channelHFET transistor Q2. The digital electrical input Q_(B) (which issupplied from the optical flip-flop 1203B in this example) is suppliedas an input to the gate terminals of both the n-channel HFET transistorQ3 and the p-channel HFET transistor Q4. In this configuration, then-channel HFET transistor Q2 is activated and operates as a turn-ontransistor for the thyristor 1400 when the binary level of the digitalelectrical input Q_(A) is high. When activated, the n-channel HFETtransistor Q2 supplies electron current to the n-channel injector.Similarly, the n-channel HFET transistor Q3 is activated and operates asa turn-on transistor for the thyristor 1400 when the binary level of thedigital electrical input Q_(B) is high. When activated, the n-channelHFET transistor Q3 supplies hole current to the p-channel injector. Thecurrents injected by both the n-channel HFET transistors Q2 and Q3 areconfigured such that the thyristor 1400 operates in its ON state whereit conducts current vertically between the anode terminal and thecathode terminal of the thyristor 1400. The p-channel HFET transistor Q1is activated and operates as a turn-off transistor for the thyristor1400 when the binary level of the digital electrical input Q_(A) is low.When activated, the p-channel HFET transistor Q1 draws electron currentfrom (i.e., supplies hole current to) the n-channel injector such thatthe thyristor 1400 operates in its OFF state where there is minimalcurrent conduction vertically between the anode and the cathode terminalof the thyristor 1400. Similarly, the p-channel HFET transistor Q4 isactivated and operates as a turn-off transistor for the thyristor 1400when the binary level of the digital electrical input Q_(B) is low. Whenactivated, the p-channel HFET transistor Q4 draws hole current from(i.e., supplies electron current to) the p-channel injector such thatthe thyristor 1400 operates in its OFF state. Thus, when both of thedigital electrical signals Q_(A) and Q_(B) are high, the thyristor 1400operates in its ON state. When either one (but not both) of the digitalelectrical signals Q_(A) and Q_(B) are high, the thyristor 1400 operatesin its OFF state. When either one (and possibly both) of the digitalelectrical signals Q_(A) and Q_(B) are low, the thyristor 1400 operatesin its OFF state.

The ON and OFF states of the thyristor 1400 generates the digitalelectrical Reset signal at the anode terminal of the thyristor 1400 anda complementary Reset signal (of opposite polarity) at the cathodeterminal of the thyristor 1400. In this configuration, the binary levelsof the digital electrical Reset signal for the ON and OFF states of thethyristor 1400 corresponds to the logical conjunction (the Boolean ANDfunction) of two digital electrical inputs Q_(A) and Q_(B) according tothe table of FIG. 12D.

FIG. 14B illustrates an embodiment of an AND gate 1205′ with opticalinputs suitable for use in the circuit of FIG. 12. The AND gate 1205′ isbasic digital logic gate produces a digital electrical output signal (inthis example, the electrical Reset signal) that is the logicalconjunction of two digital optical inputs (in this example, the digitaloptical signals Q_(A) and Q_(B)) according to the table of FIG. 12D. TheAND gate 1205′ includes a four terminal vertical thyristor (N region-Pregion-N region-P region) 1400′ with a split load. A first load element1403A is coupled between the positive voltage supply (V_(cc)) and theanode terminal of the thyristor 1400′. A second load element 1403B iscoupled between the cathode terminal of the thyristor 1400′ and thenegative voltage supply (ground). The thyristor 1400′ also has ap-channel injector terminal and an n-channel injector terminal.

A p-channel HFET transistor Q1 is coupled between the positive voltagesupply (V_(cc)) and the n-channel injector terminal of the thyristor1400′. The source terminal and gate terminal of the p-channel HFETtransistor Q1 are connected to the positive voltage supply (V_(cc)). Thedrain terminal of the p-channel HFET transistor Q1 is coupled to then-channel injector terminal of the thyristor 1400′. An n-channel HFETtransistor Q2 is coupled between the negative voltage supply (ground)and the p-channel injector terminal of the thyristor 1400′. The sourceterminal and gate terminal of the n-channel HFET transistor Q2 areconnected to the negative voltage supply (ground). The drain terminal ofthe n-channel HFET transistor Q2 is coupled to the p-channel injectorterminal of the thyristor 1400′.

Waveguides (not shown) supply both the digital optical inputs Q_(A) andQ_(B) to the waveguide region of the thyristor 1400′. In thisconfiguration, the digital optical inputs Q_(A) and Q_(B) are absorbedby the thyristor 1400′ in order to introduce majority charge carriersinto the thyristor 1400′. When both the digital optical inputs Q_(A) andQ_(B) are in the ON state, such absorbed charge is sufficient to operatethe thyristor 1400′ in its ON state where it conducts current verticallybetween anode terminal and the cathode terminal of the thyristor 1400′.The p-channel HFET transistor Q1 and the n-channel HFET transistor Q2operate as a turn-off transistors for the thyristor 1400′ when eitherone (but not both) of the digital optical inputs Q_(A) and Q_(B) are inthe ON state such that the thyristor 1400 operates in its OFF statewhere there is minimal current conduction vertically between the anodeand the cathode terminal of the thyristor 1400′. In the ON state of thethyristor 1400′, the current that conducts vertically between anodeterminal and the cathode terminal of the thyristor 1400′ generates thedigital Reset output at the anode terminal of the thyristor 1400′ andthe complementary Reset output (of opposite polarity) at the cathodeterminal of the thyristor 1400′. The bias configuration of the p-channelHFET transistor Q1 and the n-channel HFET transistor Q2 also operatesthe thyristor 1400′ in its OFF state when both the digital opticalsignals Q_(A) and Q_(B) are in the OFF state. The ON and OFF states ofthe thyristor 1400′ generates the digital Reset output at the anodeterminal of the thyristor 1400′ and the complementary Reset output (ofopposite polarity) at the cathode terminal of the thyristor 1400′. Inthis configuration, the binary levels of the digital Reset output forthe ON and OFF states of the thyristor 1400′ corresponds to the logicalconjunction (the Boolean AND function) of two digital electrical inputsQ_(A) and Q_(B) according to the table of FIG. 12D.

Optical Charge Pump

FIG. 15A illustrates an embodiment of an optical charge pump and filtercircuit 1105 for use in the circuit of FIG. 11. The optical charge pumpand filter circuit 1105 operates as a switched current source and loopfilter where positive and negative current pulses are injected into twocorresponding loop filter stages, where one loop filter stage outputs anelectrical control output (control D signal) that is used to introduce apositive change in frequency to the optical clock signal 1109A and thecorresponding electrical clock signal 1109B generated by theoptoelectronic oscillator 1101, and where the other loop filter stageoutputs an electrical control output (control C signal) that is used tointroduce a negative change in frequency to the optical clock signal1109A and the corresponding electrical clock signal 1109B generated bythe optoelectronic oscillator 1101.

The optical charge pump and filter circuit 1105 includes two fourterminal vertical thyristor (N region-P region-N region-P region) 1500and 1504 each with a split load. For the thyristor 1500, a first loadelement 1501A is coupled between the positive voltage supply (V_(cc))and the anode terminal of the thyristor 1500. A second load element1501B and a capacitor 1503 are coupled in parallel between the cathodeterminal of the thyristor 1500 and the negative voltage supply (ground).The parallel arrangement of the second load element 1501B and thecapacitor 1503 coupled between the cathode terminal of the thyristor1500 and the negative voltage supply (ground) functions as a loop filterstage that outputs an electrical control output (control D signal) thatis used to introduce a positive change in frequency to the optical clocksignal 1109A and the corresponding electrical clock signal 1109Bgenerated by the optoelectronic oscillator 1101. The thyristor 1500 alsohas a p-channel injector terminal and an n-channel injector terminal. Ann-channel HFET transistor Q1 is coupled between the positive voltagesupply (V_(cc)) and the n-channel injector terminal of the thyristor1500. The drain terminal of the n-channel HFET transistor Q1 isconnected to the positive voltage supply (V_(cc)). The source terminalof the n-channel HFET transistor Q1 is coupled to the n-channel injectorterminal of the thyristor 1500. An n-channel HFET transistor Q2 iscoupled between the negative voltage supply (ground) and the p-channelinjector terminal of the thyristor 1500. The source terminal of then-channel HFET transistor Q2 is connected to the negative voltage supply(ground). The drain terminal of the n-channel HFET transistor Q2 iscoupled to the p-channel injector terminal of the thyristor 1500. Awaveguide (not shown) supplies the digital optical input Q_(A) as inputto the waveguide region of the thyristor 1500. The digital electricalinput Q_(B) is supplied as input to the gate terminals of the n-channelHFET transistors Q1 and Q2.

For the thyristor 1504, a first load element 1505A and a capacitor 1506are coupled in parallel between the positive voltage supply (V_(cc)) andthe anode terminal of the thyristor 1504. The parallel arrangement ofthe first load element 1505A and the capacitor 1506 coupled between thepositive voltage supply and the anode terminal of the thyristor 1504functions as a loop filter stage that outputs an electrical controloutput (control C signal) that is used to introduce a negative change infrequency to the optical clock signal 1109A and the correspondingelectrical clock signal 1109B generated by the optoelectronic oscillator1101. A second load element 1505B is coupled between the cathodeterminal of the thyristor 1504 and the negative voltage supply (ground).The thyristor 1504 also has a p-channel injector terminal and ann-channel injector terminal. An n-channel HFET transistor Q3 is coupledbetween the positive voltage supply (V_(cc)) and the n-channel injectorterminal of the thyristor 1504. The drain terminal of the n-channel HFETtransistor Q3 is connected to the positive voltage supply (V_(cc)). Thesource terminal of the n-channel HFET transistor Q3 is coupled to then-channel injector terminal of the thyristor 1504. An n-channel HFETtransistor Q4 is coupled between the negative voltage supply (ground)and the p-channel injector terminal of the thyristor 1504. The sourceterminal of the n-channel HFET transistor Q4 is connected to thenegative voltage supply (ground). The drain terminal of the n-channelHFET transistor Q4 is coupled to the p-channel injector terminal of thethyristor 1504. A waveguide (not shown) supplies the digital opticalsignal Q_(B) as input to the waveguide region of the thyristor 1504. Thedigital electrical input Q_(A) is supplied as input to the gateterminals of the n-channel HFET transistors Q3 and Q4.

In this configuration, the ON level of the digital optical signal Q_(A)input (and the corresponding high level of the digital electrical signalQ_(A) input) represents the phase difference between the recovereddigital optical signal 1111 and the clock signal 1109A (where therecovered digital optical signal 1111 is ON and the clock signal 1109Ais OFF). Similarly, the ON level of the digital optical signal Q_(B)input (and the corresponding high level of the digital electrical signalQ_(B) input) represents the phase difference between the clock signal1109A and the recovered digital optical signal 1111 (where the clocksignal 1109A is ON and the recovered digital optical signal 1111 isOFF).

The thyristor 1500 absorbs the digital optical signal Q_(A) to introducemajority charge carriers in the thyristor 1500. The digital electricalsignal Q_(B) supplied as input to the gate terminal of the n-channelHFET transistor Q1 selectively activates (or deactivates) thedrain-source current path of the n-channel HFET transistor Q1. When thedigital electrical signal Q_(B) is high, the drain-source current pathof the n-channel HFET transistor Q1 is activated to drain electrons fromthe n-channel injector of the thyristor 1500. When the digitalelectrical signal Q_(B) is low, the drain-source current path of then-channel HFET transistor Q1 is deactivated such that there is minimalconduction from the n-channel injector of the thyristor 1500. Similarly,the digital electrical signal Q_(B) supplied as input to the gateterminal of the n-channel HFET transistor Q2 selectively activates (ordeactivates) the drain-source current path of the n-channel HFETtransistor Q2. When the digital electrical input Q_(B) is high, thedrain-source current path of the n-channel HFET transistor Q2 isactivated to drain holes from the p-channel injector of the thyristor1500. When the digital electrical signal Q_(B) is low, the drain-sourcecurrent path of the n-channel HFET transistor Q2 is deactivated suchthat there is minimal conduction from the p-channel injector of thethyristor 1500. Thus, with the digital electrical signal Q_(B) beinghigh, the n-channel HFET transistors Q1 and Q2 are activated to drainmajority carriers from the thyristor 1500 and counteract theintroduction of majority carriers into the thyristor as a result ofabsorption of the digital optical signal Q_(A). In this configuration,the thyristor 1500 switches ON to provide positive current that flows tothe RC load provided by the parallel load 1501B and capacitor 1503 onlywhen the digital optical signal Q_(A) is ON and the digital electricalsignal Q_(B) is low (i.e., the corresponding digital optical signalQ_(B) is OFF). With the thyristor 1500 in the ON state, the voltagepotential of the cathode terminal of the thyristor 1500 increases andthe RC load provided by the parallel load 1501B and capacitor 1503operates as a filter stage that integrates such increases in the voltagepotential of the cathode terminal over time. The result of suchintegration is an increasing voltage signal at the cathode terminalwhose change over time tracks the phase offset of the digital opticalsignal Q_(A) relative to the digital optical signal Q_(B) over time. Anexample of the increasing voltage signal generated at the cathodeterminal of the thyristor 1500 is shown in FIG. 15B. The cathodeterminal voltage signal of the thyristor 1500 is used as a Control Dsignal that is output and supplied to the optoelectronic oscillator 1101in order to introduce a positive change in frequency to the opticalclock signal 1109A and the corresponding electrical clock signal 1109Bgenerated by the optoelectronic oscillator 1101.

The thyristor 1504 absorbs the digital optical signal Q_(B) to introducemajority charge carriers in the thyristor 1504. The digital electricalsignal Q_(A) supplied as input to the gate terminal of the n-channelHFET transistor Q3 selectively activates (or deactivates) thedrain-source current path of the n-channel HFET transistor Q3. When thedigital electrical signal Q_(A) is high, the drain-source current pathof the n-channel HFET transistor Q3 is activated to drain electrons fromthe n-channel injector of the thyristor 1504. When the digitalelectrical signal Q_(A) is low, the drain-source current path of then-channel HFET transistor Q3 is deactivated such that there is minimalconduction from the n-channel injector of the thyristor 1504. Similarly,the digital electrical signal Q_(A) supplied as input to the gateterminal of the n-channel HFET transistor Q4 selectively activates (ordeactivates) the drain-source current path of the n-channel HFETtransistor Q4. When the digital electrical signal Q_(A) is high, thedrain-source current path of the n-channel HFET transistor Q4 isactivated to drain holes from the p-channel injector of the thyristor1504. When the digital electrical signal Q_(A) is low, the drain-sourcecurrent path of the n-channel HFET transistor Q4 is deactivated suchthat there is minimal conduction from the p-channel injector of thethyristor 1504. Thus, with the digital electrical signal Q_(A) beinghigh, the n-channel HFET transistors Q3 and Q4 are activated to drainmajority carriers from the thyristor 1504 and counteract theintroduction of majority carriers into the thyristor as a result ofabsorption of the digital optical signal Q_(B). In this configuration,the thyristor 1504 switches ON to provide negative current that flowsout of the RC load provided by the parallel load 1505A and capacitor1506 only when the digital optical signal Q_(B) is ON and the digitalelectrical signal Q_(A) is low (i.e., the corresponding digital opticalsignal Q_(A) is OFF). With the thyristor 1504 in the ON state, thevoltage potential of the anode terminal of the thyristor 1504 decreasesand the RC load provided by the parallel load 1505A and capacitor 1506operates as a filter stage that integrates such decreases in the voltagepotential of the anode terminal over time. The result of suchintegration is a decreasing voltage signal at the anode terminal whosechange over time tracks the phase offset of the digital optical signalQ_(B) relative to the digital optical signal Q_(A) over time. An exampleof the decreasing voltage signal generated at the anode terminal of thethyristor 1504 is shown in FIG. 15C. The anode terminal voltage signalof the thyristor 1504 is used as a Control C signal that is output andsupplied to the optoelectronic oscillator 1101 in order to introduce anegative change to the frequency of the optical clock signal 1109A andthe corresponding electrical clock signal 1109B generated by theoptoelectronic oscillator 1101.

In this configuration, the control signals C and D dynamically adjustthe frequency and phase of the optical clock signal 1109A and thecorresponding electrical clock signal 1109B generated by theoptoelectronic oscillator 1101 such that it matches the frequency andphase of the symbol clock embedded in the recovered optical signal whenit achieved the phase/frequency locked condition. This phase/frequencylocked condition is achieved when the C and D control signals becomeconstant (independent of time). This locked condition is not shown inFIGS. 15B and 15C.

Optoelectronic Oscillator

FIG. 16 shows an embodiment of an optoelectronic oscillator 1101suitable for use in the circuit of FIG. 11. The optoelectronicoscillator 1101 produces an oscillating optical output signal andcorresponding oscillating electrical output signal (i.e., an opticalclock signal 1109A and the corresponding electrical clock signal 1109Bin the example herein) whose oscillation frequency is controlled by twoelectrical input signals. One electrical input signal (i.e., the ControlC signal in the example herein) introduces a negative change to theoscillation frequency of the oscillating optical output signal (andcorresponding oscillating electrical output signal). The otherelectrical input signal (i.e., the Control D signal in the exampleherein) introduces a positive change to the oscillation frequency of theoscillating optical output signal (and corresponding oscillatingelectrical output signal).

The optoelectronic oscillator 1101 includes a four terminal verticalthyristor (N region-P region-N region-P region) 1600 with a p-channelinjector terminal and an n-channel injector terminal. An n-channel HFETtransistor Q1 is coupled between the positive voltage supply (V_(cc))and the anode terminal of the thyristor 1600. A p-channel HFETtransistor Q2 is coupled between the positive voltage supply (V_(cc))and the n-channel injector terminal of the thyristor 1500. The sourceterminal of p-channel HFET transistor Q2 is connected to the positivevoltage supply (V_(cc)). The drain terminal of the p-channel HFETtransistor Q2 is coupled to the n-channel injector terminal. A p-channelHFET transistor Q3 is coupled between the negative voltage supply(ground) and the p-channel injector terminal of the thyristor 1600. Thedrain terminal of the p-channel HFET transistor Q3 is connected to thenegative voltage supply (ground). The source terminal of the p-channelHFET transistor Q3 is coupled to the p-channel injector terminal of thethyristor 1600. An n-channel HFET transistor Q4 is coupled between thecathode terminal of the thyristor 1500 and the negative voltage supply(ground).

A feedback waveguide 1603 is coupled to the waveguide region of thethyristor 1600. The waveguide region of the thyristor 1600 acts as anoptical pulse regenerator that emits an output optical pulse in responseto a detected input optical pulse. An electrical output pulsecorresponding the output optical pulse is generated at the cathodeterminal of the thyristor 1600. The feedback waveguide structure 1603splits the output optical pulse into two parts. One part is output fromthe feedback waveguide structure 1603 to form a train of optical pulsesthat defines the output optical clock signal. The other part is guidedby the feedback waveguide 1603 such that it returns back to thethyristor 1600 as an input optical pulse to produce another outputoptical pulse. Such optical pulse regeneration can be initiated by anelectrical start-up pulse signal that is supplied to the n-channelinjector of the thyristor as shown. The corresponding electrical pulsetrain generated at the cathode terminal of the thyristor 1600 is outputas the electrical clock signal. A semiconductor optical amplifier 1605can be integral to the feedback waveguide 1603 and configured to amplifythe optical pulse signal guided by the feedback waveguide. Suchamplification can compensate for losses in splitting off the opticalclock signal from the circuit.

The pulse width of the output optical pulse (and corresponding outputelectrical pulse) is given by:

t _(pulse) =t _(intrinsic) +t _(trigger).  (5)

The parameter t_(intrinsic) is the intrinsic time delay of the device,which is based upon fabrication and growth parameters of the device (andis typically on the order of 1 to 4 picoseconds). The parametert_(trigger) is related to the effective area of the thyristor 1600,threshold charge density, and the charging current as follows:

A*σ _(trigger) =Q _(trigger) =I _(trigger) *t _(trigger).  (6)

The parameter A is the effective area of thyristor 1600. The parameterσ_(trigger) is the threshold charge density. The parameter Q_(trigger)is the threshold charge. The parameter I_(trigger) is the charging ortrigger current.

The trigger current I_(trigger) relates to the optically inducedphotocurrent and the bias current as follows:

I _(trigger) =I _(input) −I _(BiasN) −I _(BiasP).  (7)

The parameter I_(input) is the optically induced photocurrent producedby absorption of the input optical pulse. I_(BiasN) is the electroncurrent drawn from the n-channel injector of the thyristor 1600.I_(BiasP) is the hole current drawn from the p-channel injector of thethyristor 1600.

The optically induced photocurrent I_(input) is given by:

I _(input)=η_(i) *P _(input).  (8)

The parameter η_(i) represents the quantum efficiency of the thyristor1600. The parameter P_(input) is the power of input optical pulse.

Solving equation (6) for the parameter t_(trigger) using equations (7)and (8) for the parameter I_(trigger) yields:

$\begin{matrix}{{{t_{trigger} = {Q_{trigger}/I_{trigger}}};}{and}} & (9) \\{t_{trigger} = {\frac{A*\sigma_{trigger}}{( {\eta_{i}*P_{input}} ) - I_{BiasN} - I_{BiasP}}.}} & (10)\end{matrix}$

From inspection of equations (10), (9) and (5), if the optical power ofinput optical pulse is held constant (P_(input) is constant), anincrease in the bias current I_(BiasN) leads to a decrease inI_(trigger), an increase in t_(trigger), and an increase in t_(pulse).The increase in t_(pulse) decreases the frequency of the output opticalpulse train emitted from the thyristor 1600 that forms the optical clocksignal and the matching frequency of the corresponding output electricalpulse train generated at the cathode terminal of the thyristor 1600 thatforms the electrical clock signal. In contrast, a decrease in the biascurrent I_(BiasP) leads to an increase in I_(trigger), a decrease int_(trigger), and a decrease in t_(pulse). The decrease in t_(pulse)increases the frequency of the output optical pulse train emitted fromthe thyristor 1600 that forms the optical clock signal and the matchingfrequency of the corresponding output electrical pulse train generatedat the cathode terminal of the thyristor 1600 that forms the electricalclock signal.

Moreover, from the inspection of equations (10), (9) and (5), if thebias currents I_(BiasN) and I_(BiasP) are held constant, a decrease inthe optical power of the input optical pulse (decrease in P_(input))leads to a decrease in I_(trigger), an increase in t_(trigger), and anincrease in t_(pulse). The increase in t_(pulse) decreases the frequencyof the output optical pulse train emitted from the thyristor 1600 thatforms the optical clock signal and the matching frequency of thecorresponding output electrical pulse train generated at the cathodeterminal of the thyristor 1600 that forms the electrical clock signal.In contrast, an increase in the optical power of the input optical pulse(increase in P_(input)) leads to an increase in I_(trigger) a decreasein t_(trigger), and a decrease in t_(pulse). The decrease in t_(pulse)increases the frequency of the output optical pulse train emitted fromthe thyristor 1600 that forms the optical clock signal and the matchingfrequency of the corresponding output electrical pulse train generatedat the cathode terminal of the thyristor 1600 that forms the electricalclock signal.

These relationships are exploited to control the frequency of the outputoptical pulse train emitted from the thyristor 1600 that forms theoptical clock signal and the corresponding output electrical pulse traingenerated at the cathode terminal of the thyristor 1600 that forms theelectrical clock signal.

Specifically, the C control signal (which can be generated by theoptical charge pump 1105 as described in the example above) is suppliedas input to the gate terminals of the HFET transistors Q1 and Q2, andthe D control signal (which can be generated by the optical charge pump1105 as described in the example above) is supplied as input to the gateterminals of the HFET transistors Q3 and Q4. The C control signal assupplied to the gate terminal of the n-channel HFET transistor Q1controls the variable source-drain resistance of the transistor Q1. Adecrease in the C control signal increases the source-drain resistanceof the transistor Q1 and thus decreases the ON state voltage drop acrossthe thyristor 1600, thereby decreasing the power level of the outputoptical pulse. This decrease in power level results in a decrease in theoptical power of the input optical pulse (decrease in P_(input)) that isreturned to the thyristor 1600. The C control signal as supplied to thegate terminal of the p-channel HFET transistor Q2 controls the variablesource-drain resistance of the transistor Q2. A decrease in the Ccontrol signal decreases the source-drain resistance of the transistorQ2 and thus increases the bias current I_(BiasN). Thus, a decrease inthe C control signal results in a decrease in the optical power of theinput optical pulse (decrease in P_(input)) as well as an increase inthe bias current I_(BiasN). Both of these conditions decrease thefrequency of the output optical pulse train emitted from the thyristor1600 that forms the optical clock signal and the matching frequency ofthe corresponding output electrical pulse train generated at the cathodeterminal of the thyristor 1600 that forms the electrical clock signal.

The D control signal as supplied to the gate terminal of the n-channelHFET transistor Q4 controls the variable source-drain resistance of thetransistor Q4. An increase in the D control signal lowers thesource-drain resistance of the transistor Q4 and thus increases the ONstate voltage drop across the thyristor 1600, thereby increasing thepower level of the output optical pulse. This increase in power levelresults in an increase in the optical power of the input optical pulse(increase in P_(input)) that is returned to the thyristor 1600. The Dcontrol signal as supplied to the gate terminal of the p-channel HFETtransistor Q3 controls the variable source-drain resistance of thetransistor Q3. An increase in the D control signal increases thesource-drain resistance of the transistor Q3 and thus decreases the biascurrent I_(BiasP). Thus, an increase in the D control signal results inan increase in the optical power of the input optical pulse (increase inP_(input)) as well as a decrease in the bias current I_(BiasP). Both ofthese conditions increase the frequency of the output optical pulsetrain emitted from the thyristor 1600 that forms the optical clocksignal and the matching frequency of the corresponding output electricalpulse train generated at the cathode terminal of the thyristor 1600 thatforms the electrical clock signal.

In this configuration, the frequency and phase of the optical clocksignal and the corresponding electrical clock signal are adjusteddynamically by decreases in the control signal C and increases in thecontrol signal D.

POET Building Blocks

The electronic circuit components (such as the HFET transistors andelectrical thyristors) as well as the optoelectronic circuit components(such as the optical hybrid couplers and optical thyristors) of thecircuits as described herein can be implemented in one or moreintegrated circuits based on technology (referred to by the Applicant as“Planar Optoelectronic Technology” or “POET”). POET provides for therealization of a variety of devices (optoelectronic devices, logiccircuits and/or signal processing circuits) utilizing inversionquantum-well channel device structures as described in detail in U.S.Pat. No. 6,031,243; U.S. patent application Ser. No. 09/556,285, filedon Apr. 24, 2000; U.S. patent application Ser. No. 09/798,316, filed onMar. 2, 2001; International Application No. PCT/US02/06802 filed on Mar.4, 2002; U.S. patent application Ser. No. 08/949,504, filed on Oct. 14,1997, U.S. patent application Ser. No. 10/200,967, filed on Jul. 23,2002; U.S. application Ser. No. 09/710,217, filed on Nov. 10, 2000; U.S.Patent Application No. 60/376,238, filed on Apr. 26, 2002; U.S. patentapplication Ser. No. 10/323,390, filed on Dec. 19, 2002; U.S. patentapplication Ser. No. 10/280,892, filed on Oct. 25, 2002; U.S. patentapplication Ser. No. 10/323,390, filed on Dec. 19, 2002; U.S. patentapplication Ser. No. 10/323,513, filed on Dec. 19, 2002; U.S. patentapplication Ser. No. 10/323,389, filed on Dec. 19, 2002; U.S. patentapplication Ser. No. 10/323,388, filed on Dec. 19, 2002; U.S. patentapplication Ser. No. 10/340,942, filed on Jan. 13, 2003; InternationalPatent Application No. PCT/US12/51265, filed on Aug. 17, 2012; U.S.patent application Ser. No. 13/921,311, filed on Jun. 19, 2013; and U.S.patent application Ser. No. 14/222,841, filed on Mar. 24, 2014; all ofwhich are hereby incorporated by reference in their entireties.

With these structures, a fabrication sequence can be used to make thedevices on a common substrate. In other words, n type and p typecontacts, critical etches, etc. can be used to realize all of thesedevices simultaneously on a common substrate. The essential features ofthis device structure include 1) an n-type modulation doped quantum wellinterface and a p-type modulation doped quantum well interface, 2)self-aligned n-type and p-type channel contacts formed by ionimplantation, 3) n-type metal contacts to the n-type ion implants andthe bottom n-type layer structure, and 4) p-type metal contacts to thep-type ion implants and the top p-type layer structure. The activedevice structures can be realized with group III-V materials. Such groupIII-V materials can include gallium (Ga) and arsenic (As) (for galliumarsenide layer structures) as well as aluminum (Al) and indium (In), andthus can include GaAs, AlGaAs, and InGaAs semiconductor alloys.Alternatively, such group III-V materials can include gallium (Ga) andnitrogen (N) (for gallium nitride layer structures) as well as aluminum(Al) and indium (In), and thus can include GaN, AlGaN, and InGaNsemiconductor alloys.

POET can be used to construct a variety of optoelectronic devices. POETcan also be used to construct a variety of high performance transistordevices, such as complementary n-channel and p-channel HFET unipolartransistor devices as well as n-type and p-type HBT bipolar transistordevices.

Turning now to FIG. 17, the device structure of the present applicationincludes an optional bottom dielectric distributed Bragg reflector (DBR)mirror 1703 formed on substrate 1701. The bottom DBR mirror 1703 can beformed by depositing pairs of semiconductor or dielectric materials withdifferent refractive indices. When two materials with differentrefractive indices are placed together to form a junction, light will bereflected at the junction. The amount of light reflected at one suchboundary is small. However, if multiple junctions/layer pairs arestacked periodically with each layer having a quarter-wave (λ/4) opticalthickness, the reflections from each of the boundaries will be added inphase to produce a large amount of reflected light (e.g., a largereflection coefficient) at the particular center wavelength λ_(C).Deposited upon the bottom DBR mirror 1703 or upon the substrate 1701 forthe case where the bottom DBR mirror 1703 is omitted) is the activedevice structure suitable for realizing complementary heterostructurefield-effect transistor (HFET) devices. The first of these complementaryHFET devices is a p-channel HFET formed from a p-type modulation dopedquantum well (QW) structure 1711 with an n-type gate region (i.e.,n-type ohmic contact layer 1705 and n-type layer(s) 1707) below thep-type modulation doped QW structure 1711. An undoped spacer layer 1709is disposed between the p-type modulation doped quantum well (QW)structure 1711 and the underlying n-type layer(s) 1707. One or morespacer layers 1713 are disposed above the p-type modulation doped QWstructure 1711. The second of these complementary HFET devices is ann-channel HFET formed by an n-type modulation doped QW structure 1715with a p-type gate region (i.e., p-type layer(s) 1719 and p-type ohmiccontact 1721) disposed above the n-type modulation doped QW structure1715. An undoped spacer layer 1717 is disposed between the n-typemodulation doped QW structure 1715 and the overlying p-type layer(s)1719. The layers encompassing the spacer layer 1713 and the n-typemodulation doped QW structure 1715 forms the collector region of thep-channel HFET. Similarly, the layers encompassing the spacer layer 1713and the p-type modulation doped QW structure 1711 forms the collectorregion of the n-channel HFET. Such collector regions are analogous tothe substrate region of a MOSFET device as is well known. Therefore, anon-inverted n-channel HFET device can be stacked upon an invertedp-channel HFET device as part of the active device structure.

The active device layer structure begins with n-type ohmic contactlayer(s) 1705 which enables the formation of ohmic contacts thereto.Deposited on layer 1705 are one or more n-type layers 1707 and anundoped spacer layer 1709 which serve electrically as part of the gateof the p-channel HFET device and optically as a part of the lowerwaveguide cladding of the active device structure. Deposited on layer17089 is the p-type modulation doped QW structure 1711 that defines ap-type charge sheet offset from one or more QWs (which may be formedfrom strained or unstrained heterojunction materials) by an undopedspacer layer. The p-type charge sheet is formed first below the undopedspacer and the one or more QWs of the p-type modulation doped QWstructure 1711. All of the layers grown thus far form the p-channel HFETdevice with the gate ohmic contact on the bottom. Deposited on thep-type modulation doped QW structure 1711 is one or more spacer layers1713.

Deposited on the spacer layer(s) 1713 is the n-type modulation doped QWstructure 1715. The n-type modulation doped QW structure 1715 defines ann-type charge sheet offset from one or more QWs by an undoped spacerlayer. The n-type charge sheet is formed last above the undoped spacerand the one or more QWs of the n-type modulation doped QW structure1715.

Deposited on the n-type modulation doped QW structure 1715 is an undopedspacer layer 1717 and one or more p-type layers 1719 which can serveelectrically as part of the gate of the n-channel HFET and optically aspart of the upper waveguide cladding of the device. The p-type layers1719 can include two sheets of planar doping of highly doped p-materialseparated by a lightly doped layer of p-material. These p-type layersare offset from the n-type modulation doped quantum well structure 1715by the undoped spacer layer 1717. In this configuration, the top chargesheet achieves low gate contact resistance and the bottom charge sheetdefines the capacitance of the n-channel HFET with respect to the n-typemodulation doped QW structure 1715. Deposited on p-type layer(s) 1719 isone or more p-type ohmic contact layer(s) 1721, which enables theformation of ohmic contacts thereto.

For the n-channel HFET device, a gate terminal electrode of then-channel HFET device interfaces to the top p-type ohmic contactlayer(s) 1721. A source terminal electrode and a drain terminalelectrode of the n-channel HFET device are operably coupled to oppositesides of a QW channel region realized in the n-type modulation doped QWstructure 1715. One or more collector terminal electrodes can beoperably coupled to the p-type modulation doped QW structure 1711.

For the p-channel HFET device, a gate terminal electrode of thep-channel HFET device interfaces to the bottom n-type ohmic contactlayer 1705. A source terminal electrode and a drain terminal electrodeare operably coupled to opposite sides of a QW channel region realizedin the p-type modulation doped QW structure 1711. The layer structure ofthe p-channel HFET device can be patterned and etched down to form amesa at spacer layer 1713 with a collector (or back-gate) electrodeformed on such mesa.

It may be beneficial that the depth of the back-gate mesa be uniform forthe p-channel HFET devices formed on the substrate. Attempting toachieve this uniform depth by timed etching is difficult due to etchnon-uniformity and thermal variations. A solution can be found byintroducing an extremely thin layer (e.g. 10 A) of AlAs during theepitaxial growth at the level of the back-gate mesa of the p-channelHFET devices. This AlAs layer can act as an etch stop layer duringetching in order to provide a uniform depth for the back-gate mesas ofthe p-channel HFET devices formed on the substrate. In one embodiment, adry etch can be performed to above and within 600 A of the thin AlAslayer. Then a wet etch (preferably employing citric acid and H₂O₂ can beperformed. This etch process stops naturally at the AlAs layer (i.e.,the AlAs of 10 A is extremely resistant to this etching process). Thus aplanar surface is established just at the back gate level which has theaccuracy of the MBE growth.

Both the n-channel HFET device and the p-channel HFET device are fieldeffect transistors where current flows as a two-dimensional gas throughthe QW channel region with contacts on either side of the channelregion. The basic transistor action is the modulation of the QW channelconductance by a modulated electric field that is perpendicular to theQW channel. The modulated electric field modulates the QW channelconductance by controlling an inversion layer (i.e., a two-dimensionalelectron gas for the n-channel HFET device or a two-dimensional hole gasfor the p-channel HFET) as a function of gate voltage relative to sourcevoltage.

For the n-channel HFET device, the QW channel conductance is turned onby biasing the gate terminal electrode and the source terminal electrodeat voltages where the P/N junction of the gate and source regions isforward biased with minimal gate conduction and an inversion layer ofelectron gas is created in the QW channel of the n-type modulation dopedquantum well structure 1715 between the source terminal electrode andthe drain terminal electrode. In this configuration, the source terminalelectrode is the terminal electrode from which the electron carriersenter the QW channel of the n-type modulation doped quantum wellstructure 1715, the drain terminal electrode is the terminal electrodewhere the electron carriers leave the device, and the gate terminalelectrode is the control terminal for the device.

The p-channel HFET device operates in a similar manner to the n-channelHFET device with the current direction and voltage polarities reversedwith respect to those of the n-channel HFET device. For the p-channelHFET device, the QW channel conductance is turned on by biasing the gateterminal electrode and the source terminal electrode at a voltage wherethe P/N junction of the source and gate regions is forward-biased withminimal gate conduction and an inversion layer of hole gas is created inthe QW channel of the p-type modulation doped quantum well structure1711 between the source terminal electrode and the drain terminalelectrode. In this configuration, the source terminal electrode is theterminal from which the hole carriers enter the QW channel of the p-typemodulation doped quantum well structure 1711, the drain terminalelectrode is the terminal where the hole carriers leave the device, andthe gate terminal electrode is the control terminal for the device.

The device structure of the present application can also be configuredto realize bipolar inversion channel field-effect transistors (BICFETs)with either an n-type modulation doped quantum well inversion channelbase region (n-channel base BICFET) or a p-type modulation doped quantumwell inversion channel base region (p-channel base BICFET).

For the n-channel base BICFET device, an emitter terminal electrode ofthe n-channel base BICFET device interfaces to the top p-type ohmiccontact layer(s) 1721 of the active device structure. A base terminalelectrode of the n-channel base BICFET device is operably coupled to theQW channel region realized in the n-type modulation doped QW structure1715. A collector terminal electrode of the n-channel base BICFET deviceis operably coupled to the p-type modulation doped QW structure 1711.The n-channel base BICFET device is a bipolar junction type transistorwhich can be operated in an active mode by applying a forward bias tothe PN junction of the emitter and base regions while applying a reversebias to the PN junction of the base and collector regions, which causesholes to be injected from the emitter terminal electrode to thecollector terminal electrode. Because the holes are positive carriers,their injection contributes to current flowing out of the collectorterminal electrode as well as current flowing into the emitter terminalelectrode. The bias conditions also cause electrons to be injected fromthe base to the emitter, which contributes to current flowing out of thebase terminal electrode as well as the current flowing into the emitterterminal electrode.

The p-channel base BICFET device is similar in construction to thep-channel HFET device with the following adaptations. An emitterterminal electrode of the p-channel base BICFET device, which isanalogous to the gate terminal electrode of the p-channel HFET device,interfaces to the bottom n-type ohmic contact layer(s) 1721 of theactive device structure. A base terminal electrode of the p-channel baseBICFET device, which is analogous to the source or drain electrode ofthe p-channel HFET device, is operably coupled to the QW channel regionrealized in the p-type modulation doped QW structure 1711. A collectorterminal electrode of the p-channel base BICFET device, which isanalogous to the collector terminal electrode of the p-channel HFETdevice, is operably coupled to the spacer layer 1713. The p-channel baseBICFET device is a bipolar junction type transistor which can beoperated in an active mode by applying a forward bias to the PN junctionof the emitter and base regions while applying a reverse bias to the PNjunction of the base and collector regions, which causes electrons to beinjected from the emitter terminal electrode to the collector terminalelectrode. Because the electrons are negative carriers, their injectioncontributes to current flowing into the collector terminal electrode aswell as current flowing out of the emitter terminal electrode. The biasconditions also cause holes to be injected from the base to the emitter,which contributes to current flowing into the base terminal electrode aswell as the current flowing out of the emitter terminal electrode.

The active device structure of the present application can also beconfigured to realize a variety of electrical and optoelectronicthyristor devices having a vertical P-N-P-N thyristor structure. Theupper p-type region (i.e., the first P) of the vertical P-N-P-Nthyristor structure is formed by the p-type layers 1719, 1721 of theactive device structure. The upper n-type region (i.e., the first N) ofthe vertical P-N-P-N thyristor structure is formed from the n-typemodulation doped QW structure 1715 of the active device structure. Thelower p-type region (i.e., the second P) of the vertical P-N-P-Nthyristor structure is formed from the p-type modulation doped QWstructure 1711 of the active device structure. The lower n-type region(i.e., the second N) of the vertical P-N-P-N thyristor structure isformed by the bottom n-type layers 1705, 1707 of the active devicestructure.

180-Degree Optical Hybrid Coupler

FIGS. 18A-18C illustrate a configuration of a 180-degree optical hybridcoupler that can be made utilizing the layer structure of FIG. 17, whichincludes two zig-zag active waveguide structures 1801, 1803 integratedon the substrate 1701 and optically coupled to one another byevanescent-wave coupling over a gap region G. The zig-zag waveguidestructure 1801 is defined by a rib waveguide 1805 that forms a zig-zagpath. Similarly, the zig-zag waveguide structure 1803 is defined by arib waveguide 1807 that forms a zig-zag path. The optical mode thattravels through each respective rib waveguide is strongly confinedwithin the respective rib waveguide by internal reflection at thereflective interfaces of the rib waveguide. Specifically, cladding forguiding the optical mode 1809 in the rib waveguide 1807 is provided bythe top DBR mirror 1830 and the bottom DBR mirror 1703 as best shown inthe cross-section of FIG. 18B. Lateral confinement of the optical mode1809 in the waveguide 1807 is provided by refractive index changes atthe sidewalls 1811, 1813 that define the outer boundary of the waveguide18007 (FIGS. 18A and 18B), at n-type ion implants 1816 adjacent the toprib sidewalls 1811, 1813 (FIG. 18B), at the corner sidewalls, and at theinterface of the top mirror 1830 that covers the waveguide 1807. Similarstructure is used guiding the optical mode 1815 in the rib waveguide1805.

In the coupling region, the waveguides 1805 and 1807 include straightsections that extend parallel to one another and closely-spaced from oneanother by the gap region G. In the straight section of the waveguide1805, vertical confinement of the optical modes 1815, 1809 in thewaveguides 1805, 1807 can be aided by the top mirror 1830 formed tocover the top and sidewalls of the waveguides 1805, 1807 as shown.Lateral confinement of the optical mode 1815 is provided by i) arefractive index change at the periphery of the implant region 1817under the metal of the top control electrode 1819 as shown in FIG. 18C,and ii) a refractive index change at the periphery of the implant region1821 in the coupling region (gap G) as evident from FIG. 18C. In thestraight section of the waveguide 1807, lateral confinement of theoptical mode 1809 is provided by i) a refractive index change at theperiphery of the implant region 1823 under the metal of the top controlelectrode 1825 as shown in FIG. 18C, and ii) a refractive index changeat the periphery of the implant region 1821 in the coupling region (gapG) as evident from FIG. 18C.

The width (W) of the waveguides 1805, 1807 can be less than 2 μm, andpossibly 1 μm or less. The width of the gap region G (i.e., the spacingbetween the waveguides 1805, 1807) can be less than 2 μm, and possiblyon the order of 1 μm or less.

The zig-zag active waveguide structure 1801 includes the top controlterminal electrode 1819 that is electrically coupled to the top p-typeohmic contact layer 1107, a second control terminal electrode 1827 thatis electrically coupled to the n-type modulation doped QW structure 1715via an n-type ion implant 1816, and a bottom electrode 1829 that iselectrically coupled to the bottom n-type contact layer 1705 as bestshown in FIG. 18C.

The zig-zag active waveguide structure 1803 includes a top controlterminal electrode 1825 that is electrically coupled to the top p-typeohmic contact layer 1107, a second control terminal electrode 1831 thatis electrically coupled to the n-type modulation doped QW structure 1715via an n-type ion implant 1816, and a bottom electrode 1833 that iselectrically coupled to the bottom n-type contact layer 1705 as bestshown in FIG. 18C. Note that the implant regions 1817, 1821, 1823 canlocally shift the band gap in the underlying n-type modulation dopedquantum well structure 1715. This bandgap shift can prohibit chargetransfer in the QWs of the n-type modulation doped QW structure 1715across the gap region G between the adjacent waveguides 1805, 1807. Forthe waveguide 1805, voltage signals applied to the top control electrode1819 can overcome this effect to allow charge to enter (or exit) fromthe QWs of the n-type modulation doped QW structure 1715 via thecorresponding second control electrode 1827 as desired. For thewaveguide 1807, voltage signals applied to the top control electrode1525 can overcome this effect to allow charge to enter (or exit) fromthe QWs of the n-type modulation doped QW structure 1715 via thecorresponding second control electrode 1831 as desired. It is alsocontemplated that additional process steps, such as etching away the topp+ contact layers 1721 and possibly additional layers thereunder in thegap region G between the adjacent waveguides 1805 and 1807 can beperformed in order to prevent any charge transfer across the gap regionG between the adjacent waveguides 1805 and 1807.

The 180-degree hybrid coupler of FIGS. 18A to 18C includes two inputwaveguides (In0/In1) and two output waveguides (Out0/Out1). The inputwaveguide In0 is defined by one end of the waveguide 1805. The outputwaveguide Out0 is defined by the other end of the zig-zag waveguide1805. The input waveguide In1 is defined by one end of the waveguide1807. The output waveguide Out0 is defined by the other end of thezig-zag waveguide 1807. The two input waveguides In0, In1 receive twoseparate input optical signals S0 and S1 as shown. The length of thecoupling region of the two waveguides 1805 and 1807 (i.e., the straightsections that extend parallel to one another and closely-spaced from oneanother by the gap region G) is configured such that the evanescentcoupling over the coupling region of the two waveguides 1805 and 1807mixes the two input optical signals S0 and S1 to produce two outputsignals that propagate out from the Out0 and Out1 waveguides where thepower of the two input optical signals S0, S1 is split evenly (50:50split for each input signal) in each one of two output signals.

The power transfer ratio between the two input waveguides (In0/In1) maybe written as a function of Δβ, which is the propagation constantdifference due to the phase mismatch introduced by the charge differencein the two waveguides. The equation is given as:

$\begin{matrix}{T = {( \frac{\pi}{2} )^{2}\sin \; c^{2}\{ {\frac{1}{2}\lbrack {1 + ( \frac{\Delta \; \beta \; L_{o}}{\pi} )^{2}} \rbrack}^{1/2} \}}} & (11)\end{matrix}$

where L_(o)=π/2ζ is the fixed coupling length, and ζ is the couplingcoefficient.

For Δβ=0, 100% power is transferred in L_(o), and for ΔβL_(o)=√3π thereis no power transferred. For a 50:50 split where the power of the twoinput optical signals S0, S1 is split evenly in each one of two outputsignals, the values of Δβ would be used for a fixed L_(o) according toEqn. (11).

The structures of FIGS. 18A-18C can also be adapted to provide thespecific phase modulators as described above—π/4 for FIGS. 7A and 9A,and 3π/8 for FIG. 9A. The phase modulator is the simplest of devices. Astraight waveguide is injected along one or both sides with charge toproduce Δβ which corresponds to the desired phase change uponpropagating a distance L_(o), which is selected according to Eqn. (11)above.

NHFET Phototransistor

FIGS. 19A and 19B show an exemplary embodiment of an n-channel HFET(NHFET) phototransistor device 1900. A straight passive rib waveguidesection 1901 operates to passively guide light (optical mode 1903 ofFIG. 19B) into the active waveguide region of the n-channel HFETphototransistor device 1900. The straight passive rib waveguide section1901 can employ a top DBR mirror (labeled as 1930 in FIG. 19B), whichcan be realized by pairs of semiconductor or dielectric materials withdifferent refractive indices) that operates as cladding to provideguiding of the optical mode 1903 between the top DBR mirror and thebottom DBR mirror 1703 formed on the substrate 1701. The lateralconfinement of the optical mode 1903 within the passive rib waveguidesection 1901 can be provided by the index change associated withvertical sidewalls 1902 of the passive rib waveguide section 1901 andpossibly n-type ion implants in the top layers 1721, 1719, 1717 (labeledas 1905 in FIG. 19B). The n-type implants of the passive rib waveguidesection 1901 can introduce impurity free vacancy disordering into theadjacent waveguide core region when subjected to rapid thermalannealing. The bandgap of the disordered waveguide core region isincreased locally to substantially reduce absorption and associatedoptical loss. The lateral confinement of the optical mode 1903 can alsobe supported by covering the sidewalls 1902 with the top DBR mirror. Thetop DBR mirror structure can also extend in a continuous manner to formcladding over the active waveguide region of the n-channel HFETphototransistor device 1900 as is evident from the cross-section of FIG.19B. In this configuration, the top DBR mirror structure 1930 operatesas cladding for the optical mode 1903 received by the active waveguideregion of the n-channel HFET phototransistor device 1900.

The active waveguide region of the n-channel HFET phototransistor device1900 is defined by a rib waveguide formed in the top layers 1721, 1719,1717 that is aligned (both vertically and laterally) with the passiverib waveguide section 1901 as evident from FIG. 19A. The lateralconfinement of the optical mode 1903 within the active waveguide regionof the n-channel HFET phototransistor device 1900 can be provided by theindex change associated with vertical sidewalls 1907 of the rib formedin the top layers 1721, 1719, 1717 and possibly n-type ion implants 1905in the top p-type layers 1721, 1719, 1717 as shown in FIG. 19B. Then-type implants 1905 can introduce impurity free vacancy disorderinginto the adjacent waveguide core region when subjected to rapid thermalannealing. The bandgap of the disordered waveguide core region isincreased locally to substantially reduce absorption and associatedoptical loss. The lateral confinement of the optical mode 1903 can alsobe supported by covering the sidewalls 1907 with the top DBR mirror1230.

A source terminal electrode and a drain terminal electrode are operablycoupled to opposite sides of an elongate QW channel(s) realized in then-type modulation doped QW structure 1715 of the active waveguide regionof the n-channel HFET phototransistor device 1900. A collector terminalelectrode can be coupled to the p-type modulation doped QW structure1711 of the active waveguide region of n-channel HFET phototransistordevice 1900.

Specifically, the layer structure of the n-channel HFET phototransistordevice 1900 is patterned and etched to form the rib of the activewaveguide region that extends to opposed elongate intermediate mesas1909, 1911 in the spacer layer 1717 above the n-type modulation doped QWstructure 1715. The layer structure is also patterned and etched to forma vertical sidewall 1913 that defines the end of the active waveguideregion of the n-channel HFET phototransistor device 1900 opposite thepassive rib waveguide section 1901. The vertical sidewall 1913 extendsdown to an elongate intermediate mesa 1915 in the spacer layer 1713above the p-type modulation doped QW structure 1711.

N-type donor ions can be implanted through the mesas 1909, 1911 to formn-type ion implant regions 1917 that create the self-aligned n-typecontacts to the n-type modulation doped quantum well structure 1715 thatforms the QW channel(s) of the active waveguide region of the n-channelHFET phototransistor device 1900. P-type donor ions can be implantedthrough the mesa 1915 to form a p-type ion implant region that creates aself-aligned p-type contact to the p-type modulation doped quantum wellstructure 1711 of the active waveguide region of the n-channel HFETphototransistor device 1900. Deposition of a rapid thermal anneal (RTA)oxide and subsequent RTA operations can be carried out to activate theimplant regions.

The metal that defines the Source and Drain terminal electrodes isdeposited and patterned on the mesas 1909 and 1911 in contact with then-type ion implant regions 1917 in order to contact the n-typemodulation doped quantum well structure 1715 that forms the QWchannel(s) of the active waveguide region of the n-channel HFETphototransistor device 1900. The metal that defines the Collectorterminal electrode, is deposited and patterned on the mesa 1915 incontact with the p-type ion implant region in order to contact thep-type modulation doped quantum well structure of the active waveguideregion of the n-channel HFET phototransistor device 1900. The resultantstructure can be heated to treat the metals of the source, drain andcollector electrodes as desired.

The n-channel HFET phototransistor device 1900 is a field effecttransistor where current flows as a two-dimensional gas through the QWchannel region of the n-type modulation doped quantum well structure1715 of the active waveguide region with the Source and Drain terminalson either side of the QW channel region. The basic transistor action isthe modulation of the QW channel conductance by an inversion layer(i.e., a two-dimensional electron gas) that is produced by theabsorption of the optical mode 1903 propagating within the activewaveguide region of the n-channel HFET phototransistor device 1900.Specifically, the QW channel conductance is controlled by the absorptionof the optical mode 1903 propagating within the active waveguide regionof the n-channel HFET phototransistor device 1900, which produces aninversion layer of electron gas in the QW channel of the n-typemodulation doped quantum well structure 1715 between the Source terminalelectrode and the Drain terminal electrode. In this configuration, theSource terminal electrode is the terminal electrode from which theelectron carriers enter the QW channel of the n-type modulation dopedquantum well structure 1715, and the Drain terminal electrode is theterminal electrode where the electron carriers leave the device.

PHFET Phototransistor

FIGS. 20A and 20B show an exemplary embodiment of a p-channel HFET(PHFET) phototransistor device 2000. A straight passive rib waveguidesection 2001 operates to passively guide light (optical mode 2003 ofFIG. 20B) into the active waveguide region of the p-channel HFETphototransistor device 2000. The straight passive rib waveguide section2001 can employ a top DBR mirror (labeled as 2030 in FIG. 20B), whichcan be realized by pairs of semiconductor or dielectric materials withdifferent refractive indices) that operates as cladding to provideguiding of the optical mode 2003 between the top DBR mirror and thebottom DBR mirror 1703 formed on the substrate 1701. The lateralconfinement of the optical mode 2003 within the passive rib waveguidesection 2001 can be provided by the index change associated withvertical sidewalls 2002 of the passive rib waveguide section 2001 andpossibly n-type ion implants in the top layers 1721, 1719, 1717 (labeledas 2005 in FIG. 19B). The n-type implants of the passive rib waveguidesection 2001 can introduce impurity free vacancy disordering into theadjacent waveguide core region when subjected to rapid thermalannealing. The bandgap of the disordered waveguide core region isincreased locally to substantially reduce absorption and associatedoptical loss. The lateral confinement of the optical mode 2003 can alsobe supported by covering the sidewalls 2002 with the top DBR mirror. Thetop DBR mirror structure can also extend in a continuous manner to formcladding over the active waveguide region of the p-channel HFETphototransistor device 2000 as is evident from the cross-section of FIG.20B. In this configuration, the top DBR mirror structure 2030 operatesas cladding for the optical mode 2003 received by the active waveguideregion of the p-channel HFET phototransistor device 2000.

The active waveguide region of the p-channel HFET phototransistor device2000 is defined by a rib waveguide formed in layers 1721, 1719, 1717,1715, and 1713 that is aligned (both vertically and laterally) with thepassive rib waveguide section 2001 as evident from FIG. 20A. The lateralconfinement of the optical mode 2003 within the active waveguide regionof the p-channel HFET phototransistor device 2000 can be provided by theindex change associated with vertical sidewalls 2007 of the rib formedin the layers 1721, 1719, 1717, 1715, and 1713 and possibly n-type ionimplants 2005 in the top p-type layers 1721, 1719, 1717 as shown in FIG.20B. The n-type implants 2005 can introduce impurity free vacancydisordering into the adjacent waveguide core region when subjected torapid thermal annealing. The bandgap of the disordered waveguide coreregion is increased locally to substantially reduce absorption andassociated optical loss. The lateral confinement of the optical mode2003 can also be supported by covering the sidewalls 2007 with the topDBR mirror 2030.

A source terminal electrode and a drain terminal electrode are operablycoupled to opposite sides of an elongate QW channel(s) realized in thep-type modulation doped QW structure 171 of the active waveguide regionof the p-channel HFET phototransistor device 2000. A collector terminalelectrode can be coupled to the n-type modulation doped QW structure1715 of the active waveguide region of p-channel HFET phototransistordevice 2000.

Specifically, the layer structure of the p-channel HFET phototransistordevice 2000 is patterned and etched to form the rib of the activewaveguide region that extends to opposed elongate intermediate mesas2009, 2011 in the spacer layer 1713 above the p-type modulation doped QWstructure 1711. The layer structure is also patterned and etched to forma vertical sidewall 2013 that defines the end of the active waveguideregion of the p-channel HFET phototransistor device 2000 opposite thepassive rib waveguide section 2001. The vertical sidewall 2013 extendsdown to an elongate intermediate mesa 2015 in the spacer layer 1717above the n-type modulation doped QW structure 1715.

P-type donor ions can be implanted through the mesas 2009, 2011 to formp-type ion implant regions 2017 that create the self-aligned p-typecontacts to the p-type modulation doped quantum well structure 1711 thatforms the QW channel(s) of the active waveguide region of the p-channelHFET phototransistor device 2000. N-type donor ions can be implantedthrough the mesa 2015 to form an n-type ion implant region that createsa self-aligned n-type contact to the n-type modulation doped quantumwell structure 1715 of the active waveguide region of the p-channel HFETphototransistor device 2000. Deposition of a rapid thermal anneal (RTA)oxide and subsequent RTA operations can be carried out to activate theimplant regions.

The metal that defines the Source and Drain terminal electrodes isdeposited and patterned on the mesas 2009 and 2011 in contact with thep-type ion implant regions 2017 in order to contact the p-typemodulation doped quantum well structure 1711 that forms the QWchannel(s) of the active waveguide region of the p-channel HFETphototransistor device 2000. The metal that defines the Collectorterminal electrode, is deposited and patterned on the mesa 2015 incontact with the n-type ion implant region in order to contact then-type modulation doped quantum well structure of the active waveguideregion of the p-channel HFET phototransistor device 2000. The resultantstructure can be heated to treat the metals of the source, drain andcollector electrodes as desired.

The p-channel HFET phototransistor device 2000 is a field effecttransistor where current flows as a two-dimensional gas through the QWchannel region of the p-type modulation doped quantum well structure1711 of the active waveguide region with the Source and Drain terminalson either side of the QW channel region. The basic transistor action isthe modulation of the QW channel conductance by an inversion layer(i.e., a two-dimensional hole gas) that is produced by the absorption ofthe optical mode 2003 propagating within the active waveguide region ofthe p-channel HFET phototransistor device 2000. Specifically, the QWchannel conductance is controlled by the absorption of the optical mode2003 propagating within the active waveguide region of the p-channelHFET phototransistor device 2000, which produces an inversion layer ofhole gas in the QW channel of the p-type modulation doped quantum wellstructure 1711 between the Source terminal electrode and the Drainterminal electrode. In this configuration, the Source terminal electrodeis the terminal from which the hole carriers enter the QW channel of thep-type modulation doped quantum well structure 1711, and the drainterminal electrode is the terminal where the hole carriers leave thedevice.

Note that the n-channel and p-channel HFET phototransistors will havesimilar optical responses since the absorption characteristics aresimilar and the transmission of both holes and electrons is required inboth. In the n-channel HFET phototransistor, the optical mode will beguided by both sets of quantum wells; whereas, in the p-channeln-channel HFET phototransistor, the top quantum well will be removed soonly the bottom quantum well will provide guidance of the optical mode.One difference is the polarity of voltage required for operation. Then-channel HFET phototransistor operates with a positive drain-sourcevoltage so that the Source terminal electrode of the n-channel HFETphototransistor may be grounded and the light increases the channelcurrent flowing from a positive circuit node to ground. In contrast, thep-channel HFET phototransistor operates with a negative drain-sourcevoltage so that the Source terminal electrode of the p-channel HFETphototransistor may be electrically coupled to a positive supply voltageand the absorbed light increases the current flowing from the fixedpositive rail voltage to a less positive circuit node. Because of thisdifference in reference voltages, the n-channel and p-channel HFETphototransistors are ideal for steering photocurrent either into or outof an intermediate circuit node which operates at voltages between thepositive and negative supply voltages (such as the n-channel injectorterminal or p-channel injector terminal of a thyristor as describedherein). This is the configuration of the coherent balanced receiver.

The n-channel and p-channel HFET phototransistors have advantages over asimple diode detector. More specifically, a diode detector producesphotocurrent characterized by low current (high resistance) and highcapacitance. Therefore, a transimpedance amplifier is necessary toamplify the output of the diode detector in order to increase thebandwidth. However, the n-channel and p-channel HFET phototransistorshave an internal transconductance (g_(m)=dV/dI) and hence producesvoltage output directly without the need for a transimpedance amplifier.

Thyristor-Based Optoelectronic Oscillator

FIGS. 21A-21D show an embodiment of an optical thyristor 2100 formedfrom the epitaxial device structure of FIG. 17 that can be configuredfor use as the optoelectronic oscillator. The optical thyristor 2100includes a thyristor closed-path waveguide resonator 2101 spaced from asection of a first zig-zag waveguide structure 2109A by a gap region2113A as shown in FIGS. 21A and 21B. The first zig-zag waveguidestructure 2109A is optically coupled to the resonator 2101 byevanescent-wave coupling over the gap region 2113A. The opticalthyristor 2100 can also include an optional second zig-zag waveguidestructure 2109B that is spaced from the thyristor closed-path waveguideresonator 2101 by a gap region 2113B. The second zig-zag waveguidestructure 2109B is optically coupled to the resonator 2101 byevanescent-wave coupling over the gap region 2113B.

The resonator 2101 defines a resonant cavity waveguide 2102 that followsa closed path that is generally rectangular in shape. The optical pathlength of the resonant cavity waveguide 2102 is tuned to the particularwavelength of the optical mode signal that is to propagate in theresonant cavity waveguide 2102. Specifically, the length of therectangular closed path of the resonant cavity waveguide 2102 is givenas 2(L₁+L₂) and the L₁ and L₂ parameters are selected to conform to thefollowing:

$\begin{matrix}{{2( {L_{1} + L_{2}} )} = \frac{2\pi \; m\; \lambda_{c}}{n_{eff}}} & (12)\end{matrix}$

-   -   where L₁ and L₂ are the effective lengths of the opposed sides        of the active resonant cavity waveguide 2102;    -   m is an integer greater than zero;    -   λ_(C) is the center wavelength of the optical mode that is to        propagate in the resonant cavity waveguide 2102; and    -   n_(eff) is the effective refractive index of the resonant cavity        waveguide 2102.

The width (W) of the resonant cavity waveguide 2102 can be less than 2μm, and possibly 1 μm or less. The width of the gap region 2113A (i.e.,the spacing between the resonant cavity waveguide 2102 and the zig-zagwaveguide 2109A) can be less than 2 μm, and possibly on the order of 1μm or less. The width of the gap region 2113B (i.e., the spacing betweenthe resonant cavity waveguide 2102 and the zig-zag waveguide 2109B) canbe less than 2 μm, and possibly on the order of 1 μm or less.

The optical mode circulates around the resonant cavity waveguide 2102and is strongly confined within the resonant cavity waveguide 2102 byinternal reflection at the reflective interfaces of the resonant cavitywaveguide 2102. The zig-zag waveguide 2109A defines a passive ribwaveguide that forms a zig-zag path. The optical mode is stronglyconfined within the zig-zag waveguide 2109A by internal reflection atthe reflective interfaces of the zig-zag waveguide 2109A. The zig-zagwaveguide 2109B defines a passive rib waveguide that forms a zig-zagpath. The optical mode is strongly confined within the zig-zag waveguide2109B by internal reflection at the reflective interfaces of the zig-zagwaveguide 2109B.

The resonant cavity waveguide 2102 can be logically partitioned in foursections that are coupled to one another by corners as shown in FIG.21A. The four sections include a straight section that extends parallelto and is closely-spaced from a straight section of the first zig-zagwaveguide 2109A by the gap region 313A. This straight section isconfigured to provide evanescent coupling to (or from) the straightsection of the zig-zag waveguide 2109A for the optical mode signal thatcirculates in the resonant cavity waveguide 2102. The opposed straightsection is configured to provide evanescent coupling to (or from) thestraight section of the zig-zag waveguide 2109B for the optical modesignal that circulates in the resonant cavity waveguide 2102. All foursections of the resonant cavity waveguide 2102 are configured as activeportions that contribute to the detection (and possibly generation) ofthe optical mode signal that circulates in the resonant cavity waveguide2102. Such active portions can be configured with mesas, contactimplants and metallization for operation as a switching opticalthyristor.

In this configuration, the mesas, contact implants and metallizationprovide electrical contact to the top p-type ohmic contact layer 1721for the anode terminal electrode (or parts thereof) of the opticalthyristor as well as electrical contact to the n-type modulation dopedQW structure 1715 for an n-channel injector terminal electrode or partsthereof of the optical thyristor as well as electrical contact to thep-type modulation doped QW structure 1171 for a p-channel injectorterminal electrode or parts thereof of the optical thyristor as well aselectrical contact to the bottom n-type contact layer 1705 for a cathodeterminal electrode or parts thereof of the optical thyristor.

The zig-zag waveguide 2109A that extends along the gap region 2113A canbe configured with mesas, contact implants and metallization forelectrical contact to top p-type ohmic contact layer 1721 for a firstcoupling control terminal electrode 2115A as well as electrical contactto the n-type modulation doped QW structure 1715 for a second controlelectrode 2117A that are used to control the coupling coefficient of theevanescent-wave coupling between the resonant cavity waveguide 2102 andthe zig-zag waveguide 2109A. Bias circuitry (not shown) can beconfigured to control the coupling coefficient of the evanescent-wavecoupling. Specifically, the coupling coefficient of the evanescent-wavecoupling between two waveguides can be changed (i.e., modulated) bycontrolling the amount of charge (electrons) that fills the QW(s) of then-type modulation doped QW structure 1715 for the straight section ofthe resonant cavity waveguide 2102 that extends along the gap region2113A, which dictates the shifting of the absorption edge and index ofrefraction of the QW(s) of the n-type modulation doped QW structure 1715for this straight section of the resonant cavity waveguide 2102. Thebias circuitry can be realized by suitable transistor circuitry that canbe integrally formed on the substrate of the integrated circuit.

The zig-zag waveguide 2109B that extends along the gap region 2113B canbe configured with mesas, contact implants and metallization forelectrical contact to top p-type ohmic contact layer 1721 for a firstcoupling control terminal electrode 2115B as well as electrical contactto the n-type modulation doped QW structure 1715 for a second controlelectrode 2117B (not shown in FIG. 21A) that are used to control thecoupling coefficient of the evanescent-wave coupling between theresonant cavity waveguide 2102 and the zig-zag waveguide 2109B. Biascircuitry (not shown) can be configured to control the couplingcoefficient of the evanescent-wave coupling. Specifically, the couplingcoefficient of the evanescent-wave coupling between two waveguides canbe changed (i.e., modulated) by controlling the amount of charge(electrons) that fills the QW(s) of the n-type modulation doped QWstructure 1715 for the straight section of the resonant cavity waveguide2102 that extends along the gap region 2113B, which dictates theshifting of the absorption edge and index of refraction of the QW(s) ofthe n-type modulation doped QW structure 1715 for this straight sectionof the resonant cavity waveguide 2102. The bias circuitry can berealized by suitable transistor circuitry that can be integrally formedon the substrate of the integrated circuit.

As shown in FIG. 21C, the optical mode signal 2104 that circulatesaround the waveguide 2102 is strongly confined within the waveguide 2102by internal reflection at the reflective interfaces of the waveguide2102. Specifically, cladding for guiding the optical mode 2104 in thewaveguide 2102 can be provided by a top DBR mirror structure (not shown)and the bottom DBR mirror 1703. Lateral confinement of the optical mode2104 in the waveguide 2102 can be provided by: i) a refractive indexchange at the sidewalls that define the outer boundary of the waveguide2102, ii) a refractive index change at the periphery of n-type contactimplant regions 2121 under the n-channel injector electrode partslocated adjacent the sidewalls of the rib waveguide 2012, iii) arefractive index change at the periphery of the central implant region2123 located under the top anode electrode, and v) a refractive indexchange at the interface of the top DBR mirror that covers the sidewallsof the waveguide. The p-channel injector is located in a central wellthat is surrounded by the anode terminal. The well is defined by a mesain the spacer layer 1713 disposed above the p-type modulation doped QWstructure 1711. P-type donor ions can be implanted through this mesas toform a p-type ion implant region 2125 that creates a self-aligned p-typecontact to the p-type modulation doped quantum well structure 1711 ofthe active waveguide 2102.

As shown in FIG. 21D, the optical mode 2110A that propagates through thezig-zag waveguide 2109A is strongly confined within the zig-waveguide2109A by internal reflection at the reflective interfaces of thewaveguide 2109A. Specifically, cladding for guiding the optical mode2110A in the waveguide 2109A can be provided by the top DBR mirrorstructure (not shown) and the bottom DBR mirror 1703. Lateralconfinement of the optical mode 2110A in the waveguide 2109A can beprovided by i) a refractive index change at the sidewalls that definethe outer boundary of the waveguide 2109A, ii) a refractive index changeat n-type ion implant region 2131A that is disposed adjacent the top ribsidewalls of the waveguide 2109A (not shown), and iii) a refractiveindex change at the interface of the top mirror that covers thesidewalls of the waveguide 2109A. In the coupling region, the waveguide2109A includes a section that extends parallel to and is closely-spacedfrom a straight section of the microresonator waveguide 2102 by the gapregion 2113A. In this section of the waveguide 2109A, lateralconfinement of the optical mode 2110A is provided by a refractive indexchange at the periphery of the implant region 2127A under the metal ofthe coupling control electrode 2115A as shown, and a refractive indexchange at the periphery of the implant region 2129A in the couplingregion (gap 2113A) as evident from FIG. 18D. In the coupling section ofthe microresonator waveguide 2102, lateral confinement of the opticalmode 2104 is further provided by a refractive index change at theperiphery of the implant region 2129A in the coupling region (gapG—2113A) as evident from FIG. 18D. An n-type implant 2131A is disposedunder the control electrode 2117A for contact to the n-typemodulation-doped quantum well structure 1715 of the zig-zag waveguide2109A.

As shown in FIG. 21D, the optical mode 2110B that propagates through thezig-zag waveguide 2109B is strongly confined within the zig-waveguide2109B by internal reflection at the reflective interfaces of thewaveguide 2109B. Specifically, cladding for guiding the optical mode2110B in the waveguide 2109B can be provided by the top DBR mirrorstructure (not shown) and the bottom DBR mirror 1703. Lateralconfinement of the optical mode 2110N in the waveguide 2109B can beprovided by i) a refractive index change at the sidewalls that definethe outer boundary of the waveguide 2109B, ii) a refractive index changeat n-type ion implant region 2131B that is disposed adjacent the top ribsidewalls of the waveguide 2109A (not shown), and iii) a refractiveindex change at the interface of the top mirror that covers thesidewalls of the waveguide 2109A. In the coupling region, the waveguide2109B includes a section that extends parallel to and is closely-spacedfrom a straight section of the microresonator waveguide 2102 by the gapregion 2113B. In this section of the waveguide 2109B, lateralconfinement of the optical mode 2110B is provided by a refractive indexchange at the periphery of the implant region 2127B under the metal ofthe coupling control electrode 2115B as shown, and a refractive indexchange at the periphery of the implant region 2129B in the couplingregion (gap 2113B) as evident from FIG. 18D. In the coupling section ofthe microresonator waveguide 2102, lateral confinement of the opticalmode 2104 is further provided by a refractive index change at theperiphery of the implant region 2129B in the coupling region (gapG—2113B) as evident from FIG. 18D. An n-type implant 2131B is disposedunder the control electrode 2117B for contact to the n-typemodulation-doped quantum well structure 1715 of the zig-zag waveguide2109B. The cathode terminal of the thyristor closed-path waveguideresonator 2101 is formed outside the resonator on a mesa located in thebottom n-type contact layer 1705 as shown.

For configuration as an optoelectronic oscillator, the thyristorclosed-path waveguide resonator 2101 can be configured for digitaloptical-to-electrical conversion where the optical signal propagating inthe waveguide 2102 generates photocurrent by absorption which addselectrons to the n-type modulation doped QW structure 1715 and holes tothe p-type modulation doped QW structure 1711 such that the thyristordevice switches ON and conducts current through the device between theanode terminal electrode and the cathode terminal electrode. Suchoptoelectronic operations provide the function of detection,current-to-voltage conversion (typically provided by a transimpedanceamplifier), level shifting to obtain a ground reference and a decisioncircuit (typically realized by a comparator). Moreover, the thyristorclosed-path waveguide resonator 2101 has an advantage that it will onlyabsorb at the resonator frequency and thus can be adapted to supportdifferent wavelengths for wavelength division multiplexing applications.Moreover, the ON state current that flows between the anode terminalelectrode and the cathode terminal electrode of the thyristor resonatoris configured above the threshold for lasing I_(TH), such that photonemission will o_(cc)ur within the device structure. In principle, theresonator produces light that propagates in both the clockwise andcounterclockwise sense along the optical path of the resonant cavitywaveguide 2102. Moreover, the evanescent coupling across the gap region2113A between the resonant cavity waveguide 2102 and the zig-zagwaveguide 1209A operates on both clockwise and counterclockwise lightpropagation within the resonant cavity waveguide 2102. Similarly, theevanescent coupling across the gap region 2113B between the resonantcavity waveguide 2102 and the zig-zag waveguide 1209B operates on bothclockwise and counterclockwise light propagation within the resonantcavity waveguide 2102.

In this configuration, the input optical pulse can be supplied to oneend of the zig-zag waveguide 2019A as shown in FIG. 21A such that itpropagates through the straight section of the zig-zag waveguide 2019Aadjacent the gap region 2113A, where it couples into the waveguide 2102by evanescent coupling for propagation within the waveguide 2102 asshown by the arrows of FIG. 21B. The input pulse signal that propagateswithin the waveguide 2102 is then subject to digitaloptical-to-electrical conversion and pulse regeneration by the thyristorswitch action as described herein.

A bragg-grating reflector 2121 is optically coupled to the zig-zagwaveguide structure 210B9 opposite its output end as shown in FIG. 21A.The bragg-grating reflector 212 is a linear active waveguide deviceformed as a light reflector that builds upon the multiple directions oflight propagation in the resonant cavity waveguide 2102 of the resonatoras well as the multiple directions of evanescent coupling providedbetween the zig-zag waveguide 2109B and the resonant cavity waveguide2102 of the resonator. The bragg-grating reflector 2121 has Bragggrating 2123 (such as a first-order or third-order Bragg grating)defined throughout the length of the active waveguide device. The Bragggrating 2123 can be defined by etching into the top layers (such aslayers 1721, 1719, 1717 of layer structure of FIG. 17) as best shown inFIGS. 21E and 21F. The Bragg grating 2123 operates to reflect anyoptical modes propagating in the zig-zag waveguide 2109B in thedirection of the reflector 2121 where the wavelength of such opticalmodes coincides with the Bragg frequency of the Bragg grating 2123. Alloptical modes at other wavelengths will be passed through the tuningreflector 2121 or be absorbed. The Bragg grating 2123 can be configuredsuch that the Bragg frequency of the Bragg grating 2123 closely matchesthe desired output wavelength for the resonator.

The pulse regeneration that results from the thyristor switching actionof the resonator produces optical mode(s) that propagate clockwisewithin the resonant cavity waveguide 2102 of the resonant, which arecoupled into the zig-zag waveguide 2109B to produce optical mode(s) thatpropagate in the zig-zag waveguide 2109B to the reflector 321. Theincident optical mode(s) at wavelengths that coincide with the Braggfrequency of the Bragg grating 2123 of the reflector 2121 are reflectedback and propagate in the reverse direction within the zig-zag waveguide2109B where the mode is coupled into the resonator to produce opticalmode(s) that propagates counter-clockwise in the resonant cavitywaveguide 2102 of the resonator and generate more stimulated emission.This operation is repeated many times such that the wavelength ofdominant optical mode that propagates in the resonant cavity waveguide2102 of the resonator corresponds to the Bragg frequency of the Bragggrating 2123. Such dominant optical mode propagating counter-clockwisein the resonant cavity waveguide 2102 of the resonator is coupled to thezig-zag waveguide 2109B to produce the regenerated output optical pulsesignal (which propagates in the direction away from the tuning reflector2121 as shown in FIG. 21A). In this manner, optical modes that coincidewith the Bragg frequency of the Bragg grating 2123 of the reflector 2121make double passes through the resonant cavity waveguide 2102 of theresonator for improved stimulated emission. This operation is repeatedsuch that the wavelength of the dominant or primary optical mode thatpropagates in the resonant cavity waveguide 2102 of the resonatorcorresponds to the Bragg frequency of the Bragg grating 2123. With thisoperation, the dominant or primary optical mode that propagates in theresonant cavity waveguide 2102 of the resonator is output from thezig-zag waveguide 2109B, while optical modes that do not coincide withthe Bragg frequency of the Bragg grating 2123 are removed from theoutput pulse signal by the operation of the reflector 2121.

The Bragg grating 2123 functions as a narrow-band filter where the Braggfrequency of grating 2123 dictates the wavelength of the dominant orprimary optical mode that propagates in the resonant cavity waveguide2102 of the resonator. Such narrow-band filtering is useful for largerclosed-loop resonators where the natural mode resonances are closelyspaced from one another and thus do not provide a narrow wavelength bandfor the optical mode that propagates in the closed-loop resonator. It isalso contemplated that the Bragg frequency of the Bragg grating 2123 canbe electrically-controlled (or tuned) by controlled injection of chargethat modifies the index of the region n2 of the Bragg grating 323 asdescribed in U.S. patent application Ser. No. 14/222,841, filed on Mar.24, 2014, commonly assigned to assignees of the present invention andherein incorporated by reference in its entirety.

It is also contemplated that the optical thyristor 2100 of FIGS. 21A-21Ccan readily be configured as an optical thyristor for other circuitfunctions described herein. In one example, the optical thyristor 2100of FIGS. 21A-21C can be configured as part of the optical phase detectorof FIGS. 3A, 10E, 10H, and 10I where the output optical signal thatresults from the switching action of the optical thyristor is outputfrom output end of waveguide 2109B. In another example, the opticalthyristor 2100 of FIGS. 21A-21C can be configured as part of the opticalXOR gate of FIG. 8A where the output optical signal C that results fromthe switching action of the optical thyristor is output from output endof waveguide 2109B. In yet another example, the optical thyristor 2100of FIGS. 21A-21C can be configured as part of the optical flip-flops ofFIGS. 13A and 13B where the input optical signal (D_(A) or D_(B)) aresupplied to the one end of waveguide 2109A (similar to the input opticalpulse in FIG. 21A), and the output optical signal (Q_(A) or Q_(B)) thatresults from the switching action of the optical thyristor is outputfrom output end of waveguide 2109B. In yet another example, the opticalthyristor 2100 of FIGS. 21A-21C can be configured as part of theoptical-input AND gate of FIG. 14A where the input optical signal Q_(A)is supplied to the one end of waveguide 2109A (similar to the inputoptical pulse in FIG. 21A), and the input optical signal Q_(B) issupplied to the other end of waveguide 2109A. In still another example,the optical thyristor 2100 of FIGS. 21A-21C can be configured as part ofthe optical charge pump of FIG. 15A where the input optical signal(Q_(A) or Q_(B)) is supplied to the one end of waveguide 2109A (similarto the input optical pulse in FIG. 21A).

Polarization Diversity Systems

It is also common for PSK and QAM modulation schemes to be combined withpolarization diversity where the modulated carrier wave signal employslight with orthogonal polarizations. In such systems, the coherentoptical receiver designs as described herein can be replicateddownstream of optical elements that split the received optical signalinto multiple legs corresponding to the orthogonal polarizations andthen processes each leg to isolate the optical signal of thecorresponding polarization state for detection.

Wavelength Division Multiplexing (WDM) Systems

It is also common for PSK and QAM modulation schemes to be combined withwavelength division multiplexing where the modulated carrier wavesignals can be defined by specific wavelengths of light. In suchsystems, the coherent optical receiver designs as described herein canbe replicated downstream of optical elements that split the receivedoptical signal into multiple legs corresponding to the specific lightwavelengths and then processes each leg to isolate the optical signal ofthe corresponding wavelength for detection.

There have been described and illustrated herein several embodiments ofa coherent optical receiver and parts thereof that can be used for awide variety of communications and data processing applications. Whileparticular embodiments of the invention have been described, it is notintended that the invention be limited thereto, as it is intended thatthe invention be as broad in scope as the art will allow and that thespecification be read likewise. It will therefore be appreciated bythose skilled in the art that yet other modifications could be made tothe provided invention without deviating from its spirit and scope asclaimed.

1. An optoelectronic circuit for producing an optical clock signal, theoptoelectronic circuit comprising: an optical thyristor; a waveguidestructure that is configured to split an optical pulse produced by theoptical thyristor such that a first portion of the optical pulse isoutput as part of the optical clock signal and a second portion of theoptical pulse is guided back to the optical thyristor to produce anotheroptical pulse that is output as part of the optical clock signal; and acontrol circuitry, operably coupled to the optical thyristor, thatreceives first and second control signal inputs, wherein the controlcircuitry is configured to selectively decrease a frequency of theoptical clock signal based on the first control signal input and toselectively increase the frequency of the optical clock signal based onthe second control signal input.
 2. The optoelectronic circuit of claim1, wherein: the optical thyristor is defined by an epitaxial layerstructure that includes a bottom n-type cathode region, an intermediatep-type region formed above the bottom n-type cathode region, anintermediate n-type region formed above the intermediate p-type region,and a top p-type anode region formed above the intermediate n-typeregion, and wherein the optical thyristor further includes an anodeterminal electrically coupled to the top p-type anode region, an n-typeinjector terminal electrically coupled to the intermediate n-typeregion, a p-type injector terminal electrically coupled to theintermediate p-type region, and a cathode terminal electrically coupledto the bottom n-type cathode region.
 3. The optoelectronic circuit ofclaim 2, wherein: the control circuitry includes a first field-effecttransistor whose source-drain current path is electrically coupledbetween a positive voltage supply and the anode terminal, and whereinthe first control signal input is supplied to the first field-effecttransistor in order to control an output power of the optical pulse suchthat the frequency of the optical clock signal decreases.
 4. Theoptoelectronic circuit of claim 3, wherein: the first field-effecttransistor comprises an n-channel HFET transistor having a gate terminalthat receives the first control signal input, and wherein a decrease inthe first control signal input, decreases the output power of theoptical pulse and the frequency of the optical clock signal.
 5. Theoptoelectronic circuit of claim 2, wherein: the control circuitryincludes a second field-effect transistor whose source-drain currentpath is electrically coupled between a positive voltage supply and then-type injector terminal, and wherein the first control signal input issupplied to the second field-effect transistor in order to control abias current, thereby decreasing the frequency of the optical clocksignal.
 6. The optoelectronic circuit of claim 5, wherein: the secondfield-effect transistor comprises a p-channel HFET transistor having agate terminal that receives the first control signal input, and whereina decrease in the first control signal input increases the bias currentand decreases the frequency of the optical clock signal.
 7. Theoptoelectronic circuit of claim 2, wherein: the control circuitryincludes a third field-effect transistor whose source-drain current pathis electrically coupled between the cathode terminal and ground, andwherein the second control signal input is supplied to the thirdfield-effect transistor in order to control an output power of theoptical pulse such that the frequency of the optical clock signalincreases.
 8. The optoelectronic circuit of claim 7, wherein: the thirdfield-effect transistor comprises an n-channel HFET transistor having agate terminal that receives the second control signal input, and whereinan increase in the second control signal input increases the outputpower of the optical pulse and the frequency of the optical clocksignal.
 9. The optoelectronic circuit of claim 2, wherein: the controlcircuitry includes a fourth field-effect transistor whose source-draincurrent path is electrically coupled between the p-type injectorterminal and ground, and wherein the second control signal input issupplied to the fourth field-effect transistor in order to control abias current, thereby increasing the frequency of the optical clocksignal.
 10. The optoelectronic circuit of claim 9, wherein: the fourthfield-effect transistor comprises a p-channel HFET transistor having agate terminal that receives the second control signal input, and whereinan increase in the second control signal input decreases the biascurrent and increases the frequency of the optical clock signal.
 11. Theoptoelectronic circuit of claim 1, wherein: the waveguide structureincludes an optical amplifier device configured to amplify the secondportion of the optical pulse.
 12. The optoelectronic circuit of claim 2,wherein: the intermediate n-type and intermediate p-type regions of theepitaxial layer structure include an n-type modulation doped quantumwell (QW) structure and a p-type modulation doped QW structure,respectively.
 13. The optoelectronic circuit of claim 12, wherein: thecontrol circuitry includes at least one field effect transistor thatincludes an n-type QW channel formed by the n-type modulation doped QWstructure of the epitaxial layer structure.
 14. The optoelectroniccircuit of claim 12, wherein: the control circuitry includes at leastone field effect transistor that includes a p-type QW channel formed bythe p-type modulation doped QW structure of the epitaxial layerstructure.
 15. The optoelectronic circuit of claim 2, wherein: theepitaxial layer structure comprises group III-V materials.
 16. Theoptoelectronic circuit of claim 1, wherein: the optical thyristorproduces an electrical clock signal, and wherein a frequency of theelectrical clock signal matches the frequency of the optical clocksignal.
 17. The optoelectronic circuit of claim 1, wherein: the opticalthyristor is realized by an optical resonator including a closed pathwaveguide that supports circulating propagation of light and a waveguidestructure that is spaced from the closed path waveguide to provide foran evanescent-wave optical coupling therebetween.
 18. The optoelectroniccircuit of claim 17, wherein: the waveguide structure of the opticalresonator has one end disposed opposite an output end, and the opticalresonator further includes a reflector structure integral to the one endof the waveguide structure, and wherein the reflector structure includesa Bragg-grating.
 19. An optical phase lock loop comprising: anoptoelectronic circuit comprising: an optical thyristor; a waveguidestructure that is configured to split an optical pulse produced by theoptical thyristor such that a first portion of the optical pulse isoutput as part of an optical clock signal and a second portion of theoptical pulse is guided back to the optical thyristor to produce anotheroptical pulse that is output as part of the optical clock signal; and acontrol circuitry, operably coupled to the optical thyristor, thatreceives first and second control signal inputs, wherein the controlcircuitry is configured to selectively decrease a frequency of theoptical clock signal based on the first control signal input and toselectively increase the frequency of the optical clock signal based onthe second control signal input; an optical phase detector that measuresa phase difference between a reference optical signal and the opticalclock signal, and generates an output; and a feedback circuit that isconfigured to generate the first and second control signal inputs basedon the output of the optical phase detector for supply to theoptoelectronic circuit.
 20. The optical phase lock loop of claim 19,wherein: the feedback circuit comprises a charge pump circuit.
 21. Theoptical phase lock loop of claim 20, wherein: the charge pump circuitcomprises at least one other optical thyristor.
 22. The optical phaselock loop of claim 19, wherein: the optical phase detector comprises atleast one optical flip-flop realized by another optical thyristor. 23.The optical phase lock loop of claim 19, wherein: the optical phasedetector comprises an optical AND gate realized by another opticalthyristor.
 24. The optical phase lock loop of claim 19, wherein: theoptical phase detector comprises an AND gate realized by a thyristor.25. The optical phase lock loop of claim 19, which is configured toperform a clock recovery function.